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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 1 Designing Static CMOS Logic Circuits

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 2 Static CMOS Circuits At every point in time (except during the switching transients) each gate output is connected to either V DD orV ss via a low-resistive path. The outputs of the gates assume at all times the value of the Boolean function, implemented by the circuit (ignoring, once again, the transient effects during switching periods).

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 3 Static Complementary CMOS V DD F(In1,In2,…InN) In1 In2 InN In1 In2 InN PUN PDN PMOS only NMOS only PUN and PDN are logically dual logic networks … …

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 4 NMOS Transistors in Series/Parallel Connection Transistors can be thought as a switch controlled by its gate signal NMOS switch closes when switch control input is high

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 5 PMOS Transistors in Series/Parallel Connection

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 6 Threshold Drops V DD V DD 0 PDN 0 V DD CLCL CLCL PUN V DD 0 V DD - V Tn CLCL V DD V DD |V Tp | CLCL S DS D V GS S SD D

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 7 Complementary CMOS Logic Style

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 8 Example Gate: NAND

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 9 Example Gate: NOR

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 10 Complex CMOS Gate OUT = D + A (B + C) D A BC D A B C

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 11 Constructing a Complex Gate Logic Dual need not be Series/Parallel Dual In general, many logical dual exist, need to choose one with best characteristics Use Karnaugh-Map to find good duals Goal: find 0-cover and 1-cover with best parasitic or layout properties Maximize connections to power/ground Place critical transistors closest to output node C(a) pull-down networkSN1SN4SN2SN3DFFADBCDFABC(b) Deriving the pull-up networkhierarchically by identifyingsub-netsDAABCV DD V B(c) complete gate

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 12 Example: Carry Gate F = (ab+bc+ac) Carry c is critical Factor c out: F=(ab+c(a+b)) 0-cover is n-pd 1-cover is p-pu CC AB

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 13 Example: Carry Gate (2) Pull Down is easy Order by maximizing connections to ground and critical transistors For pull up – Might guess series dual– would guess wrong

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 14 Example: Carry Gate (3) Series/Parallel Dual 3-series transistors 2 connections to Vdd 7 floating capacitors

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 15 Example: Carry Gate (4) Pull Up from 1 cover of Kmap Get ab+ac+bc Factor c out 3 connections to Vdd 2 series transistors Co-Euler path layout Moral: Use Kmap!

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 16 Cell Design Standard Cells General purpose logic Can be synthesized Same height, varying width Datapath Cells For regular, structured designs (arithmetic) Includes some wiring in the cell Fixed height and width

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 17 Standard Cell Layout Methodology – 1980s signals Routing channel V DD GND

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 18 Standard Cell Layout Methodology – 1990s M2 No Routing channels V DD GND M3 V DD GND Mirrored Cell

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 19 Standard Cells Cell boundary N Well Cell height 12 metal tracks Metal track is approx Pitch = repetitive distance between objects Cell height is 12 pitch 2 Rails ~10 In Out V DD GND

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 20 Standard Cells In Out V DD GND InOut V DD GND With silicided diffusion With minimal diffusion routing

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 21 Standard Cells A Out V DD GND B 2-input NAND gate

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 22 Stick Diagrams Contains no dimensions Represents relative positions of transistors In Out V DD GND Inverter A Out V DD GND B NAND2

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 23 Stick Diagrams C AB X = C (A + B) B A C i j j V DD X X i GND AB C PUN PDN A B C Logic Graph

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 24 Two Versions of C (A + B) X CABABC X V DD GND V DD GND

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 25 Consistent Euler Path j V DD X X i GND AB C ABC

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 26 OAI22 Logic Graph C AB X = (A+B)(C+D) B A D V DD X X GND AB C PUN PDN C D D A B C D

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 27 Example: x = ab+cd

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 28 Properties of Complementary CMOS Gates Snapshot High noise margins: V OH andV OL are atV DD andGND, respectively. No static power consumption: There never exists a direct path betweenV DD and V SS (GND) in steady-state mode. Comparable rise and fall times: (under appropriate sizing conditions)

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 29 CMOS Properties Full rail-to-rail swing; high noise margins Logic levels not dependent upon the relative device sizes; ratioless Always a path to Vdd or Gnd in steady state; low output impedance Extremely high input resistance; nearly zero steady-state input current No direct path steady state between power and ground; no static power dissipation Propagation delay function of load capacitance and resistance of transistors

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 30 Switch Delay Model A R eq A RpRp A RpRp A RnRn CLCL A CLCL B RnRn A RpRp B RpRp A RnRn C int B RpRp A RpRp A RnRn B RnRn CLCL NAND2 INV NOR2

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 31 Input Pattern Effects on Delay Delay is dependent on the pattern of inputs Low to high transition both inputs go low –delay is 0.69 R p /2 C L one input goes low –delay is 0.69 R p C L High to low transition both inputs go high –delay is R n C L CLCL B RnRn A RpRp B RpRp A RnRn C int

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 32 Delay Dependence on Input Patterns A=B=1 0 A=1, B=1 0 A=1 0, B=1 time [ps] Voltage [V] Input Data Pattern Delay (psec) A=B= A=1, B= A= 0 1, B=1 61 A=B= A=1, B= A= 1 0, B=1 81 NMOS = 0.5 m/0.25 m PMOS = 0.75 m/0.25 m C L = 100 fF

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 33 Transistor Sizing CLCL B RnRn A RpRp B RpRp A RnRn C int B RpRp A RpRp A RnRn B RnRn CLCL

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 34 Multi-Fingered Transistors One finger Two fingers (folded) Less diffusion capacitance

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 35 Transistor Sizing a Complex CMOS Gate OUT = D + A (B + C) D A BC D A B C

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 36 Fan-In Considerations DCBA D C B A CLCL C3C3 C2C2 C1C1 Distributed RC model (Elmore delay) t pHL = 0.69 R eqn (C 1 +2C 2 +3C 3 +4C L ) Propagation delay deteriorates rapidly as a function of fan-in – quadratically in the worst case.

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 37 t p as a Function of Fan-In t pL H t p (psec) fan-in Gates with a fan-in greater than 4 should be avoided. t pH L quadratic linear tptp

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 38 t p as a Function of Fan-Out t p NOR2 t p (psec) eff. fan-out All gates have the same drive current. t p NAND2 t p INV Slope is a function of driving strength

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 39 t p as a Function of Fan-In and Fan- Out Fan-in: quadratic due to increasing resistance and capacitance Fan-out: each additional fan-out gate adds two gate capacitances to C L t p = a 1 FI + a 2 FI 2 + a 3 FO

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 40 Practical Optimization The previous arguments regarding tp raise the question – why build nor at all? Criticality is not a path– but a transition so it is usually only on rising or falling (but not both) NOR forms have bad pull-up but good pull down NAND forms have bad pull-down but good pull up Determine the critical transition(s) and design for them– using Elmore or Simulation on the appropriate edge only! Logical Effort presupposes uniform rise and fall times, so good in general, but can be beat Static Timing Analyzers nearly always get this wrong!

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 41 Fast Complex Gates: Design Technique 1 Transistor sizing as long as fan-out capacitance dominates Progressive sizing In N CLCL C3C3 C2C2 C1C1 In 1 In 2 In 3 M1 M2 M3 MN Distributed RC line M1 > M2 > M3 > … > MN (the fet closest to the output is the smallest) Can reduce delay by more than 20%; decreasing gains as technology shrinks

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 42 Fast Complex Gates: Design Technique 2 Transistor ordering C2C2 C1C1 In 1 In 2 In 3 M1 M2 M3 CLCL C2C2 C1C1 In 3 In 2 In 1 M1 M2 M3 CLCL critical path charged charged 1 delay determined by time to discharge C L, C 1 and C 2 delay determined by time to discharge C L charged discharged

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 43 Fast Complex Gates: Design Technique 3 Alternative logic structures F = ABCDEFGH

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 44 Fast Complex Gates: Design Technique 4 Isolating fan-in from fan-out using buffer insertion CLCL CLCL

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 45 Fast Complex Gates: Design Technique 5 Reducing the voltage swing linear reduction in delay also reduces power consumption But the following gate may be much slower! Large fan-in/fan-out requires use of sense amplifiers to restore the signal (memory) t pHL = 0.69 (3/4 (C L V DD )/ I DSATn ) = 0.69 (3/4 (C L V swing )/ I DSATn )

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 46 Sizing Logic Paths for Speed Frequently, input capacitance of a logic path is constrained Logic also has to drive some capacitance Example: ALU load in an Intels microprocessor is 0.5pF How do we size the ALU datapath to achieve maximum speed? We have already solved this for the inverter chain – can we generalize it for any type of logic?

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 47 Buffer Example For given N: C i+1 /C i = C i /C i-1 To find N: C i+1 /C i ~ 4 How to generalize this to any logic path? CLCL InOut 12N (in units of inv )

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 48 Logical Effort p – intrinsic delay (3kR unit C unit ) - gate parameter f(W) g – logical effort (kR unit C unit ) – gate parameter f(W) f – effective fanout Normalize everything to an inverter: g inv =1, p inv = 1 Divide everything by inv (everything is measured in unit delays inv ) Assume = 1.

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 49 Delay in a Logic Gate Gate delay: d = h + p effort delay intrinsic delay Effort delay: h = g f logical effort effective fanout = C out /C in Logical effort is a function of topology, independent of sizing Effective fanout (electrical effort) is a function of load/gate size

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 50 Logical Effort Inverter has the smallest logical effort and intrinsic delay of all static CMOS gates Logical effort of a gate presents the ratio of its input capacitance to the inverter capacitance when sized to deliver the same current Logical effort increases with the gate complexity

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 51 Logical Effort Logical effort is the ratio of input capacitance of a gate to the input capacitance of an inverter with the same output current g = 1 g = 4/3 g = 5/3

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 52 Logical Effort of Gates Fan-out (h) Normalized delay (d) t pINV t pNAND F(Fan-in) g = p = d = g = p = d =

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 53 Logical Effort of Gates Fan-out (h) Normalized delay (d) t pINV t pNAND F(Fan-in) g = 1 p = 1 d = h+1 g = 4/3 p = 2 d = (4/3)h+2

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 54 Add Branching Effort Branching effort:

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 55 Multistage Networks Stage effort: h i = g i f i Path electrical effort: F = C out /C in Path logical effort: G = g 1 g 2 …g N Branching effort: B = b 1 b 2 …b N Path effort: H = GFB Path delay D = d i = p i + h i

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 56 Optimum Effort per Stage When each stage bears the same effort: Minimum path delay Effective fanout of each stage: Stage efforts: g 1 f 1 = g 2 f 2 = … = g N f N

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 57 Optimal Number of Stages For a given load, and given input capacitance of the first gate Find optimal number of stages and optimal sizing Substitute best stage effort

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 58 Logical Effort From Sutherland, Sproull

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 59 Method of Logical Effort Compute the path effort: F = GBH Find the best number of stages N ~ log 4 F Compute the stage effort f = F 1/N Sketch the path with this number of stages Work either from either end, find sizes: C in = C out *g/f Reference: Sutherland, Sproull, Harris, Logical Effort, Morgan-Kaufmann 1999.

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 60 Example: Optimize Path Effective fanout, F = G = H = h = a = b = g = 1 f = a g = 5/3 f = b/a g = 5/3 f = c/b g = 1 f = 5/c

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 61 Example: Optimize Path g = 1 f = a g = 5/3 f = b/a g = 5/3 f = c/b g = 1 f = 5/c Effective fanout, F = 5 G = 25/9 H = 125/9 = 13.9 h = 1.93 a = 1.93 b = ha/g 2 = 2.23 c = hb/g 3 = 5g 4 /f = 2.59

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 62 Example: Optimize Path Effective fanout, H = 5 G = 25/9 F = 125/9 = 13.9 f = 1.93 a = 1.93 b = fa/g 2 = 2.23 c = fb/g 3 = 5g 4 /f = 2.59 g 1 = 1g 2 = 5/3g 3 = 5/3 g 4 = 1

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 63 Example – 8-input AND

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 64 Summary Sutherland, Sproull Harris

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EE141 © Forrest Brewer and © Digital Integrated Circuits 2nd Combinational Circuits 65 Homework 5 1. Using the Carry cell design from earlier homework, optimally size the carry propagate chain for a 16-bit adder to minimize worst case delay where Cin is driven by a 1u/0.6u inverter and Cout drives a fanout of 4 such loads. (use logic effort, show your work!) 2. For the problems below, use parameters from class for 0.5um and use 2x voltages as applicable. Chap 5: problems: 4, 7, 8, Chap 6, problems: 2, 4, 5, 7 4. Design the parity tree: c = a xor b xor c xor d in Complementary Pass Transistor Logic, insert inverters to restore the output swing – Given input drive from an inverter stage, and an inverter every 2 stages of logic, and inverter output restore, estimate the propagation time for devices using the AMI 0.5um model. New (digital) AMI model (for minimum length only!): n-channel: V T =0.77, =0.03, V sat =1.56V, k=32 A/V 2 p-channel: V T =-0.95, =0.03 V sat =2.8V, k=-16 A/V 2

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