1. High Linearity and High Efficiency Power Amplifiers in GaN HEMT Technology
Thank you for being my committee and thank you for coming to my qualify today. Let’s start. My topic of today is: High linearity and high efficiency PA design in GaN HEMT.
Shouxuan Xie
Department of Electrical and Computer Engineering,
University of California, Santa Barbara
June 30, 2003

2. Outline 1. Introduction and motivation - Why GaN HEMTs
- Objectives of the GaN HEMTs PA design
Class B for high efficiency and high linearity
- Why single-ended Class B
- Circuit design and measurement result
Identify and model nonlinear sources of GaN HEMTs
- Nonlinear gm
- Nonlinear Cgs
- Nonlinear Gds
Proposed new designs to further improve linearity
- Common drain Class B (to improve gm nonlinearity)
- Pre-linearization diode (to improve Cgs nonlinearity)
5. Problems and future works
Here is the outline. First, I will introduce the objective of the project. Then I will show you the measurement result of the single ended Class B PA that I designed last year. Then I will discuss some nonlinear sources of the GaN HEMT device and their effects on the linearity performance of the circuit. Base on that, I will show you the two new circuits that we designed in order to further improve the linearity. Finally I will show you the problems that I have right now, and the proposed future works. Some of them you already seen during Paidi’s qualify two week ago, for example the analysis of class B and the new design of common drain. But in order to make my talk logical and completely, I will mention them again. But I will go through quickly.

3. Why GaN HEMTs Advantages of GaN - High breakdown field: 3 MV/cm
Standard AlGaN/GaN HEMT structure
Advantages of GaN
- High breakdown field: 3 MV/cm
- High 2.5 x 107 cm/s
- Thermal conductivity: 3x GaAs
- Large channel charge: > 1x1013 cm-2
- Good electron mobility: >1200 cm2/V-s
This is the structure of a standard AlGaN/GaN HEMT. The advantages of the GaN HEMT are list here. Basically it has very high breakdown, high ft, and high power density at RF frequencies.
Advantages of GaN HEMTs
- High power density: 12W/mm for X-band (8-12GHz)
- High Ft (50GHz) and fmax (80GHz) for 0.25um device
- Linear I-V characteristics

4. GaN HEMT process and device structure
0.25um T-gate for 50GHz ft
Air-bridge for ground
connection of CPW
MIM capacitors
Here is the device layer structure of the GaN HEMT, and some other processing issues that we are currently using in the cleanroom here in UCSB, for example MIM capacitor, SiN passivation, and air-bridges for circuit.
SiN passivation for High RF output power
SiC substrate for high heat conducting

5. Device performance I-V Curve for 600m SG device
Linear Id-Vgs characteristic on SiC
Idss = 1
And here are the multi-finger device performance on SiC. And we got 1A per millimeter of Idss with a linear I-V characteristic like here. And we got 50 GHz of ft and 55V of breakdown voltage for the dual gate device.
RF Performance 150m DG device
Device performance summary:
Lg ~ 0.25um,
Idss ~ 1A/mm
ft ~ 55GHz (50GHz for DG)
Vbr ~ 40V (55V for DG)

6. Objectives of GaN HEMT PA design
Design RF MMIC power amplifier in GaN HEMT technology to achieve:
1. High linearity (low IMD3 distortion)
2. High efficiency
3. High output power
4. Broad bandwidth
(High linearity and high efficiency are primarily concerned here)
So base on the device structure and performance, the goal of my project is to design a RF MMIC PA with high linearity, high efficiency, high output power and wide bandwidth. And high linearity and efficiency are primarily concerned. As we know that Class A has high linearity but very poor efficiency, while Class D or E has high efficiency but very poor linearity and bandwidth, therefore, in order to get high linearity and efficiency simultaneously, class B is a good candidate.
Class A: Very high linearity and wide bandwidth; but very low efficiency (Ideal PAE 50%, feasible PAE 20-30%).
Switch mode Amplifiers (Class D, E): Very high efficiency (Ideal PAE 100%, feasible PAE %); but poor linearity and poor bandwidth.
Class B: Good efficiency (Ideal PAE 78.6%; feasible PAE 40-50% ) and good bandwidth, and potentially low distortion.

7. Push-pull Class B
So here is the schematic of a classic push pull class B power amplifier. It consists two identical devices which are working in 50% duty cycle with 180 degree phase shift. Therefore, the even harmonics are suppressed by symmetry of this structure. The bandwidth of the this configuration is then limited only by the power combiner (transformers). But at RF frequencies, we have to use baluns instead of transformers. Unfortunately, most of the existing RF baluns can not provide a wideband even harmonic short. And also the baluns will occupy a lot of expensive die area, which will increase the cost. Therefore, push pull is not suitable for RF MMIC PA.
Even harmonics are suppressed by symmetry => wide bandwidth
Half-sinusoidal current is needed at each drain. This requires an even-harmonic short. It can be achieved at HF/VHF frequencies with transformers or bandpass filters. However,
Most wideband microwave baluns can not provide effective short for even-mode. Efficiency is then poor.
They occupy a lot of expensive die area on MMIC.

8. Single-ended = push-pull
Push-pull Class B
ID1
Vin
-Vin +
-ID2
= ID
180
+Vin
-Vin
ID2
ID1
But in fact, from the linearity point of view, a single ended class B with a bandpass filter is equivalent to push pull class B. It can be shown from the Taylor expansion of the transfer function. For push pull, the even harmonic is suppressed by symmetry, and for the Single ended class B, the harmonics are filtered out by the bandpass filter. But the IM3 distortion term, which is coming from the 3rd order term here, can not be filtered out because it is in band. (So from the linearity point of view, we can say that push pull is equivalent to single ended class B because they have the same transfer function). In another word, push pull is not necessary because it does not improve the IM3 performance, or say, linearity. But there is a restriction for single ended class B in bandwidth because of the filter, which make the bandwidth less than 2 to 1.
Even harmonics suppressed by symmetry
Single-ended Class B with bandpass filter
Zero Z at 2f0 ID
Even harmonics suppressed by filter
Conclusion: From linearity point of view, push-pull and single-ended Class B with bandpass filter B are equivalent – same transfer function.
Bandwidth restriction < 2:1

9. Class B bias for high linearity
ID1
Vin +
ID2
= ID Vp
ID1
Vin +
ID2
= ID Vp
ID1
Vin +
ID2
= ID Vp
This slide shows that biasing at Class B is very critical for linearity. If the transfer function of the device is ideally linear like that, in fact, then bias exactly at pinch off point is necessary. When we bias a little bit lower, which is at Class C, however, there will be a short section during which both device are off. So the linearity of Class C is very poor. Likewise, the last one shows if bias a little bit higher, which is Class AB, there will be a short section during which both device are on, therefore the transconductance will be twice as the single one. This kink will also make Class AB nonlinear.
Ideal Class B
Bias too low: Class C
Bias too high: Class AB

10.  - section lowpass filter
Single-ended Class B Power Amplifier
Here is I designed the single ended class B PA. This is the circuit diagram of the amplifier. Dual gate device is used here because it has higher breakdown, higher maximum stable gain, and higher output resistance. At input, a lossy input matching network is used to increase the bandwidth. At the output, the output capacitance of the device Cds is absorbed into the matching network to form a pi shape lowpass filter, instead of a bandpass filter, which is also to increase the bandwidth and also efficiency.
 - section lowpass filter
Lossy input matching
Dual gate device is used since it has higher Vbr, higher MSG (smaller S12) and higher output resistance Rds
Lossy input matching network to widen the bandwidth
Cds is absorbed into output matching network (Low pass filter)

11. Measurement setup Measurements: Single tone from 4 GHz to 12 GHz;
Here is our test setup. We did single tone measurement, from 4 to 12 GHz, and two tone measurement, around 8 GHz, separated by 1MHz. We also sweep the bias voltage to get Class A, B, C and AB bias conditions.
Measurements:
Single tone from 4 GHz to 12 GHz;
Two-tone measurement at f1 = 8 GHz, f2 = GHz;
Bias sweep: Class A (Vgs = -3.1V), Class B (Vgs = -5.1V), Class C (Vgs = V) and AB (Vgs = -4.5 V).

12. Class B PA measurement results
Gain and bandwidth
Class AB
So here are the measurement results. First for Class B bias condition, the gain is about 13dB around 9GHz, and the 3dB bandwidth is 3GHz, from 7 to 10 GHz. (And the gain for class A or AB are about 6dB higher than Class B as we expected)
Class B
3 dB bandwidth for Class B: 7GHz - 10GHz

13. Class B bias @Vgs = - 5.1V Two tone performance @
Single tone f0 = 8GHz:
Saturated output power 36 dBm
PAE (saturated) ~ 34%
Here is the measurement result for class B. For the single tone at 8GHz, we got 36 dBm maximum output power, and 34% saturated PAE. Here is the two tone measurement, and the red line is the IM3 power. It can be seen that we got good IM3 performance as we expected, greater than 35 dBc for Pin below 17.5dBm.
f1,f2
Two tone
f1=8GHz, f2=8.001GHz :
2f1-f2, 2f2-f1
Good IM3 performance:
40dBc at Pin = 15 dBm
> 35 dBc for Pin < 17.5 dBm

14. Class A bias @Vgs = - 3.1V Single tone performance @ f0 = 8GHz:
Saturated output power 36 dBm
PAE (saturated) ~ 34%
This is the measurement result for class A bias condition. We got the same maximum output power and saturated PAE as class B for single tone, but for two tone, Class A shows good IM3 performance at low power levels, but becomes bad rapidly when power goes higher.
Two tone f1=8GHz, and f2=8.001GHz :
f1,f2
2f1-f2, 2f2-f1
Good IM3 performance at low power level but becomes bad rapidly at high power levels

15. IM3 suppressions of all Classes
Class B
Class A
Class C
Class AB
Psat
Here we compare the IM3 performance vs output power for all the classes. At low output power levels, both class A and Class B are linear, (larger than 36dbc for class B, larger than 45 dbc for class A). But at high output power levels, class A behaves almost the same as class B. It is also shown that both class AB and class C are not linear compared to class B. All of these confirm our analysis at the beginning.
Low output power levels (Pout < 24 dBm), Class A and Class B both exhibit good linearity (Class B > 36 dBc, Class A > 45 dBc).
Higher output power levels, Class A behaves almost the same as Class B.
Class AB and C exhibit more distortion compared to Class A and B.

16. Class B vs. Class A
PAE of single tone
IM3 suppression and PAE of two-tone
Class A
Class B
This slide compares the PAE and IM3 performance between class A and Class B only. Let’s look at here where Pout = 25 dBm, both Class A and Class B can get 35 dBc in IM3, but Class B has 25% PAE in single tone at this power level while it is only 14% for class A. Therefore it proves that class B is better than class A. And by biasing at Class B, we can get good linearity and efficiency simultaneously.
Now the next step will be how to improve linearity for the class B PA. In order to do that, first we need to know where are the nonlinearity come from. In another word, the nonlinear sources of the GaN HEMT PA.
Class B
Class A
Maintaining good IM3 suppression, Class B can get 10% PAE improvement over Class A during low distortion operation.

17. Nonlinear sources of GaN HEMT
1. gm vs. Vgs of 600um SG device
Goal: Try to investigate nonlinear sources of the GaN HEMT device and understand how they affect the linearity on circuit
Three major sources have been investigated:
1. Nonlinear gm ( or Ids -Vgs characteristic)
2. Nonlinear Cgs
3. Nonlinear Gds
So what I show you here are some device parameters of the GaN HEMT that are not linear as function of bias conditions, which are potentially the nonlinear sources of the PA. Here three major nonlinear sources are investigated, Gm, Cgs, and Gds. Now I will discuss the nonlinear effect that they bring one by one.
3. Gds vs. Vgs and Vds of 600um SG device
2. Cgs vs. Vgs of SG device
Vds=20V
Vds=15V
Vds=10V

18. Nonlinear sources of GaN HEMT
Input MN
(linear, Zs)
Cgs
Cds
Gds RL
This is the equivalent circuit diagram of the GaN HEMT PA. The nonlinear sources gm, cgs and Gds are located in different places, which will cause nonlinear effect in different ways. Now I am going to show you how they affect the overall linearity performance of the circuit.

19. Nonlinear gm Cgs Cds Gds RL Input MN (linear, Zs)
Now first let’s look at nonlinear gm, in another word, the nonlinear transfer function of the device. The I-V transfer characteristic of the device can be expressed by polynomials of Vgs like this. It is clearly known that third order intermodulation distortion will be generated since there is a cubic term of the Vgs.

20. Nonlinear gm Modeled as: This term creates IM3 distortion
So this is the real I-V curve of a GaN HEMT device. We can see that the coefficient of the cubic term is small but not zero because nonlinearity of here and here. The nonlinearity of here will cause IMD3 distortion at high output power levels, which is what we are more interested in, since at here the output power is high and hence efficiency is high. So this explains how gm will generate IM3 distortion.
Dominate at high
output power levels – more interesting Vp
Dominate at low
output power levels

21. Nonlinear Cgs - Directly effect of Cgs + Q(Vgs) Cgs Cds Gds RL
Input MN
(linear, Zs)
Cgs -
Cds +
Gds RL
Then let’s look at the effect of Cgs. In general, a nonlinear Cgs can be described by the total charge on the capacitor as a function of the voltage across it, like this. Suppose a linear signal, which only has frequency component w0, is applied to the input. (Since the voltage of the capacitor will not change immediately, initially the voltage across the capacitor will contain only w0.) This voltage will generate charges on the capacitor, which is a function of the voltage like this. It will then generate current. Then this current will produce voltage drop on the linear input MN, hence the total voltage drop on the Cgs will be changed. So if there is a cubic term of Vgs in the charge vs Vgs, it will produce 3w0 in the current, hence 3w0 component back on the voltage. In a word, the nonlinear Cgs will generate third order components at the input, then get amplified by the device.
Q(Vgs)

22. Nonlinear Cgs direct Cgs vs Vgs of GaN HEMTs on SiC If modeled as:
Anti-symmetric about V=Vc
then should be no distortion
So this is a Cgs vs Vgs curve of a real GaN HEMT device. The dots are measurement data and the line is fitted by the model. Since it is this cubic term in Q(Vgs) curve creates IM3, if we look at Cgs vs. Vgs curve, it will corresponding to the square term of Vgs. So the even order component in Cgs vs. Vgs curve will create IM3 distortion. But if we model the Cgs as TanH function of Vgs, then it is purely anti-symmetric around Vc, which means if we bias at Vc, the nonlinear Cgs will not generate third order distortion. This seems to be a good news. But it is not true, because number one, in reality, Cgs is not purely anti-symmetric due to here and if Vp is not equal to Vc. Number two is the square term of Vgs will also generate third order distortion indirectly, which I am going to show you next.
Vc  Vp
Therefore even order component of Cgs(Vgs) creates IM3 distortion
This term creates IM3 distortion

23. Nonlinear Cgs – Indirect effect
Input MN
(linear, Zs)
Cgs - +
Cds
Q(Vgs)
Gds RL
Suppose, there is no cubic term in charge Q(Vgs), then as we discuss before, for the first iteration, Vgs will only have w0 and 2w0 components, but no 3w0 component.

24. Nonlinear Cgs – Indirect effect
Input MN
(linear, Zs)
Cgs - +
Cds
Q(Vgs)
Gds RL
But during the second iteration, the Vgs who contain w0 and also 2w0 will generate 3w0 component in charge due to the square Vgs term here, hence 3w0 component in current then finally back to the voltage. This means, third order component can also be generated at the gate indirectly by the square term in Q vs. Vgs relation.

25. Nonlinear Cgs + nonlinear gm
Input MN
(linear, Zs)
Cgs
Cds
Gds
Likewise, if we consider nonlinear Cgs and nonlinear gm together, even there is no 3w0 component in the input Vgs, third order distortion will still be generated by not only the cubic term in gm directly, but also by the square term of Vgs indirectly.
direct
Indirect

26. Nonlinear Gds Gds vs. Vgs of 600um SG device Cgs Cds Rds RL Input MN
(linear, Zs)
Cgs
Cds
Rds RL
The third one is nonlinear Gds. The relation between Gds and Vgs is shown here, and it is not linear. Therefore, it will produce nonlinear components directly to the output, just like other nonlinear effect that we discuss before.
Gds vs. Vgs of 600um SG device
Vds=20V
Vds=15V
Vds=10V

27. Nonlinear Gds DC I-V curve of 600um device on SiC Short channel effect
Vgs = 0 V
Another issue associated with nonlinear Gds is that Gds is also a function of Vds. This can be seen from the IV curve. The slope here and here are different. This is due to the so-called short channel effect. When close to pinch off, electrons are not well confined in the 2-DEG channel, and can be leak through deep in the GaN buffer layer if it is not semi-insulating. Therefore need more gate voltage to pinch it off.
Vgs = -7V
Vds = 8V
Vgs = -7 V
Vds = 15V
Current through GaN buffer, need more gate voltage to pinch off

28. Vp shift due to short channel effect
1.2mm SG device DC I-V curve at different drain bias
Vds=20V
Vds=15V
This is a plot of I-V curves of SG device under different drain bias Vds. It clearly shows that the pinch off voltage is shifted from -5V to -7V when the Vds changes from 10V to 20V. How bad this shift will be?
Vp shift
Vds=10V

29. Nonlinear Cgs + Vp shiftVc V i n C g s -C 2 Vb
Cgs(Vin)
Cgs(-Vin)
It will generate third order distortion through nonlinear Cgs, which I am going to explain here. As we discussed at beginning, we need to bias the circuit exactly at Vp, which means Class B to get high linearity. Suppose the Cgs vs Vgs is pure anti-symmetric, then the even order component can be easily shown as the following here in the picture. In general, even order component of a function f(x) is equal to f(x) itself, plus f(-x), and then divided by 2. So if the Vp is shifted and we still want to bias at Vp, then it will generate second order component in Cgs, which will create third order distortion.
Now we see that how those 3 nonlinear sources create distortion. And we can see that it is complicated, (directly and also indirectly). And they also correlated to each other, which make things more complicate. DC
Even order
component
Vb=Vp=Vc
Vb<Vc
Vb>Vc
Vb>>Vc

30. Paidi’s nonlinear model
Nonlinear Gds currently is modeled by shift in Vp;
The model that we used initially is the Curtice cubic model. It consider everything together, so it is difficult to understand how each nonlinear source affect the final linearity performance the from the simulation result. In order to separate the effects of the nonlinear sources, So we need a good and simple model. So here is a new model, proposed by paidi. The idea is to model the nonlinear components by self-defined function blocks directly in ADS. The transfer function block is here, currently the gm is constant. And the nonlinear Cgs block is here, which is still by tanH. And the nonlinear Gds is only treated as shift in Vp.
Cgs is ideal tanH
I-V characteristic currently is linear

31. Further improve linearity
1. Common drain Class B to improve gm linearity
CD circuit schematic
Now I am going to describe two new circuits that we designed, which are to further improve the linearity. The first one, which is designed by paidi, is common drain class B PA instead of common source. The idea of common drain is the load resistance RL in common drain configuration will also behave as a series- series feedback resistor, which increase the linearity in gm. This is the gain of common drain, comparing the gain of common source here. So this is the linearization factor. In the condition, if RL*gm is large enough, the gain is nearly equal to one. But the problem of CD is that it is not unconditionally stable because the MSG is low for CD. Therefore, extra requirement for the source and load impedance is needed.
Linearization factor
RL also functions as series-series feedback resistor, which increase gm linearity. RL
Disadvantage -- Stability problem: Since the MSG is less, the circuit is not unconditionally stable in order to keep reasonable high efficiency. Therefore, extra requirement for the source and load impedance is needed.

32. Simulation result of CD @5GHz
Pout and PAE in single tone
Pout ~ 38dBm
Pout
Here is the simulation result of the designed common drain class B PA at 5 GHz. 38dBm maximum output power is obtained with 38% saturation PAE.
PAE
PAE(sat) ~ 38%

33. Simulation result of CD vs. CS – cont.
Two-tone simulation result of CD vs. CS
Common Drain
10 dB
In order to compare between common drain and common source, I designed another common source class B PA at 5GHz with exactly the same 1.2mm SG device. Two tone simulation result are shown here. Red one is IM3 suppression for CD, and blue one is for CS. We can see that for common drain 12dB improvement in IM3 is obtained over common source at Pout equal to 20dBm, and about 10dB improvement at pout equal to 30dBm(1W).
12 dB
Common Source: with 37.6dBm Pout and 42% PAE(sat)

34. Common Drain vs. Common Source – cont.
Simulation result of IM3 suppression at 1W total output power as a function of bias point
Class C
Common Drain
Class AB
Simulation result also shows that at least 10dB improvement for CD over CS can be obtained for all the bias conditions, which is very good and what we expect.
Common Source
Class B
Class A

35. Further improve linearity – cont.
2. Pre-linearization diode to improve Cgs linearity
C_total
The second idea to improve linearity is to add another capacitor at the gate to compensate the nonlinearity of Cgs. So if the Cgs of the device is like this, which can be modeled as TanH, and if I can add another capacitor, which behaves exactly the opposite around Vc, then the total equivalent input capacitor will be constant, and therefore there will be no distortion due to Cgs, no matter where I bias my circuit.
Cgs
C_pd Vc

36. Pre-linearization diode
Vb1=Vp=-4V
0.25umx100umx12
This idea can be easily implemented by add another HEMT device at the gate, which is shown in this circuit diagram. If the Vp of the device is -4V, and if I bias the gate of the PD at twice of the Vp, which is -8V, the Csg of the diode will behave exactly opposite to Cgs, when we changing the input signal. And this idea can be simply implemented on chip, and occupy almost no extra area in layout, which is shown here. Another good thing is that, since the gates are written by e-beam, the gate length of the diode can be varied and optimum value of Csg can be found.
Vb1=2*Vp=-8V
Can be very easily implemented on chip and occupy very small area
Gate length can be varied and optimum value can be found since write using E-beam-lithography
0.75umx100umx4

37. Simulation result of PD
IM3 simulation result the designed dual gate CS Class B with pre-linearization
At least 4dBc improvement in IMD3
With PD
So two more circuits are designed at 10 GHz, both using 1.2mm Dual Gate device, but one with the diode, and one without. At least 4dB improvement in IMD3 is obtained with the PD around 25dBm output power level. (gm nonlinearity is not included in the model currently, and the Vp shift in Cgs is small for dual gate, the improvement is only 4dBc. But we should notice that we are talking about 40dBc of IM3 suppression, which means the third order power is already very low, only 0.01% of the fundamental power)
Without PD

38. Problems and future works
!! Problem: Short channel effect for 0.25um device !!
0.25umx100um device on Sapphire
0.75umx100um device on Sapphire
Vgs=0V
Vgs=0V
Problems an future works. According all the analysis above, the main problem I have now is the short channel effect of the 0.25um gate device. Here is the DC and 80um pulse IV curve for 0.25um device, and here is for 0.7um device to compare. So we can see the extra current here is due to short channel effect, not because of gate leakage or bad GaN buffer layer. This short channel effect is significant for circuit performance, because the nonlinear effects caused by this will not be seen if bias drain voltage Vds is less than 8V or 10V.
Vgs=-10V
Vds=16V
Vgs=-7V
Nonlinear Gds will affect linearity performance directly;
It creates Vp shift, hence generate nonlinear Cgs distortion;
Increases DC bias current, hence decreases PAE;
Decreases breakdown voltage, hence decreases the output power and also PAE …

39. Short channel effect - Number of gates get doubled, hard to yield all
Currently dual gate device is used:
- Nearly no Vp shift
- Lower Gds (higher Rds)
- Higher maximum stable gain (MSG)
I-V curve of 600um DG device
Gds of 600um devices at Vds=20V
Vds =15V
Currently I am using dual gate device, which is to minimize this effect. There is almost no Vp shift from the IV curve since the Vds dropped on the first device is almost constant, and the Rds of the dual gate device is almost four times larger than that of single gate device. But the disadvantage for dual gate device are
Single gate
Vds =20V
Dual gate
- Number of gates get doubled, hard to yield all
- Little bit lower ft, and higher Vknee, hence lower PAE
- Not easy to model the nonlinear effect

40. Layouts of the new designed circuits
CD SG Class
CS SG Class
So here are the layouts of the new designed circuits that I am currently fabricating now: The common drain and common source at 5 GHz, and the common source class B with and without the PD.
CS DG Class
CS DG Class with PD

41. New device structures to improve linearity
To further improve the performance, new device structure is used. A ion doped layer is added to make the buffer more semi-insulating, which to decrease the leakage current through the buffer, hence increase Rds. To solve the problem of short channel effect, one idea suggested by Prof. Mishra is to make the ion doping layer as close as to the channel. Gate recess can also increase the aspect ratio, which may partially solve the short channel effect. But how will they affect the overall linearity performance of the circuit is unclear right now. But at least there are something that I can try. And I am thinking that what if I decrease the Al percentage in AlGaN, which can increase the breakdown, and also decrease gm.
Improve short channel effect by:
- Make the Fe doping layer closer to the channel
- Gate recess to increase aspect ratio
Add Fe doping layer to decrease leakage current through the buffer
??? Question:
How about decrease Al% in AlGaN -Increase breakdown and decrease gm?
How about P-type doping GaN buffer layer?
??? Other ideas to increase breakdown???

42. Summary Class B bias is good for high linearity and high efficiency;
Three main nonlinear sources of the GaN HEMT device have been investigated with a new idea of nonlinear model;
According to simulation, common drain class B can improve linearity by 10dB over CS, and pre-linearization diode can improve linearity by 4dB. Four more circuits are designed and being fabricated to prove them;
Short channel effect for 0.25um device has been observed. New device structure is proposed to solve the problem and better linearity performance is expected.

43. Proposed future works
1. Fabricate and measure the new designed circuits (CD and PD)
- Need to stabilize the PECVD passivation process
2. Complete the new model to understand all the nonlinear effects
- Add gm nonlinearity
- More accurate model for dual gate device
3. Further improve linearity by new device structures
- Work with Mishra’s group to improve the short channel effect
4. Publish paper and write thesis
5. New ideas on device structure and model to further increase linearity and efficiency
summer
summer
So here are the proposed future works and the time schedule.
Fall
Fall

44. Publications and references
Vamsi Paidi, Shouxuan Xie, R. Coffie, U. Mishra, M J W Rodwell, S. Long, “Simulations of High linearity and high efficiency of Class B Power Amplifiers in GaN HEMT Technology.”  Lester Eastman Conference, Aug. 2002
Shouxuan Xie, Vamsi Paidi, R. Coffie, S. Keller, S. Heikman, A. Chini, U. Mishra, S. Long, M. Rodwell, “High Linearity Class B Power Amplifiers in GaN HEMT Technology.” Topical Workshop on Power Amplifiers, Sept. 2002
Shouxuan Xie, Vamsi Paidi, R. Coffie, S. Keller, S. Heikman, A. Chini, U. Mishra, S. Long, M.J.W. Rodwell, “High linearity of Class B Power Amplifiers in GaN HEMT technology.” Microwave and Wireless Components Letters, to be published
Vamsi Paidi, Shouxuan Xie, R. Coffie, B. Moran, S. Heikman, S. Keller, A. Chini, S. P. DenBaars, U. K. Mishra, S. Long and M. J.W. Rodwell, “High Linearity and High Efficiency of Class B Power Amplifiers in GaN HEMT Technology.”  IEEE Transactions on Microwave Theory and Techniques, Vol. 51, No. 2, Feb. 2003
Finally this is a list of my publication related to this topic, and also a list of some of the references. Thank you very much!!
Other references:
K. Krishnamurthy, R. Vetury, S. Keller, U. Mishra, M. J. W. Rodwell and S. I. Long, “ Broadband GaAs MESFET and GaN HEMT Resistive Feedback Power Amplifiers.” IEEE Journal of Solid State Circuits, Vol. 35, No. 9, Sept
K. Krishnamurthy, S. Keller, C. Chen, R. Coffie, M. Rodwell, U. K. Mishra, “Dual-gate AlGaN/GaN Modulation-doped Field-effect Transistors with Cut-Off Frequencies ƒT >60 GHz”, IEEE Electron Device Letters, Vol. 21, No. 12, Dec. 2000

45. Publications and references- cont.
Solid State Radio Engineering, Herbert L. Krauss, W. Bostian, Frederick H. Raab/ Wiley, John & Sons, Nov. 1980
Raab, F.H. Maximum efficiency and output of class-F power amplifiers. IEEE Transactions on Microwave Theory and Techniques, vol.49, (no.6, pt.2), IEEE, June p
Kobayashi, H.; Hinrichs, J.M.; Asbeck, P.M. “Current-mode class-D power amplifiers for high-efficiency RF applications”. IEEE Transactions on Microwave Theory and Techniques, vol.49, (no.12), IEEE, Dec p
Eastman, L.F.; Green, B.; Smart, J.; Tilak, V.; Chumbes, E.; Hyungtak Kim; Prunty, T.; Weimann, N.; Dimitrov, R.; Ambacher, O.; Schaff, W.J.; Shealy, J.R. Power limits of polarization-induced AlGaN/GaN HEMT's. Proceedings 2000 IEEE/ Cornell Conference on High Performance Devices, Piscataway, NJ, USA: IEEE, p pp..
Wu, Y.-F.; Kapolnek, D.; Ibbetson, J.; Zhang, N.-Q.; Parikh, P.; Keller, B.P.; Mishra, U.K. “High Al-content AlGaN/GaN HEMTs on SiC substrates with very high power performance”. International Electron Devices Meeting 1999, Piscataway, NJ, USA: IEEE, p pp.
Joseph, J. Teaching design while constructing a 100-watt audio amplifier. Proceedings. Frontiers in Education 1997, 27th Annual Conference (vol.1)Pittsburgh, PA, USA, 5-8 Nov ) Champaign, IL, USA: Stipes Publishing, p vol.1. 3 vol. xxxvi+1624 pp. 3
Shealey, V.; Tilak, V.; Prunty, T.; Smart, J.A.; Green, B.; Eastman, L.F.” An AlGaN/GaN high-electron-mobility transistor with an AlN sub-buffer layer”. Journal of Physics: Condensed Matter, vol.14, (no.13), IOP Publishing, 8 April p
W. R. Curtice and M. Ettenberg, "A nonlinear GaAsFET model for use in the design of output circuits for power amplifiers," IEEE Trans of Microwave Theory Tech, vol. MTT-33, pp , Dec

46. Thank you!

51. Does Vc change? Vc Consider Cgs nonlinearity only
simulate IM3 result at 1W output power level:
Vp = -5V, Vc = -5V, without PD: 46.3dBc, with PD: 57.4dBc
Vp = -5.5V, Vc = -5V, without PD: 40.1dBc, with PD: 57.6dBc

54. GaN HEMT Model – Vp shift

56. Advantages of GaN GaN: ND = 1017 cm-3 GaN: ND = 1019 cm-31 2 3 4
Electron Velocity (107 cm/sec)
T = 300 K
3.0
Note scale change
( 10 larger)
GaN: ND = 1017 cm-3
GaN: ND = 1019 cm-3
0.5
1.5
1.0
2.0
2.5
Electric Field Strength (105 V/cm)
0.3
Ref: Gelmont et al.,
J. Applied Physics
74, August 1, 1993
GaAs
InP
InGaAs Si

58. Class B two-tone output spectrum
Medium input power 1
Low input power
Pout =18 dBm
IM3 = 39 dBc
Pout = 4 dBm
IM3 = 43 dBc
Medium input power 2
High input power
Pout = 26 dBm
IM3 = 25 dBc
Pout = 22 dBm
IM3 = 40 dBc

59. Class A two-tone output spectrum
Low input power
Medium input power 2
Pout = 10 dBm
IM3 > 50 dBc
Pout = 23 dBm
IM3 = 42 dBc
Medium input power 2
High input power
Pout = 27 dBm
IM3 = 31 dBc
Pout = 31 dBm
IM3 = 15 dBc