1. Zero-voltage transition converters The phase-shifted full bridge converter
Buck-derived full-bridge converter
Zero-voltage switching of each half-bridge section
Each half-bridge produces a square wave voltage. Phase-shifted control of converter output
A popular converter for server front-end power systems
Efficiencies of 90% to 95% regularly attained
Controller chips available

2. Issues with this converter
It’s a good converter for many applications requiring isolation. But…
Secondary-side diodes operate with zero-current switching. They require snubbing or other protection to avoid failure associated with avalanche breakdown
The resonant transitions reduce the effective duty cycle and conversion ratio. To compensate, the transformer turns ratio must be increased, leading to increased reflected load current in the primary-side elements
During the D’Ts interval when both output diodes conduct, inductor Lc stores energy (needed for ZVS to initiate the next DTs interval) and its current circulates around the primary-side elements—causing conduction loss

3. Result of analysis Basic configuration: full bridge ZVT

4. Actual waveforms, including resonant transitions

5. Issues with this converter
It’s a good converter for many applications requiring isolation. But…
Secondary-side diodes operate with zero-current switching. They require snubbing or other protection to avoid failure associated with avalanche breakdown
The resonant transitions reduce the effective duty cycle and conversion ratio. To compensate, the transformer turns ratio must be increased, leading to increased reflected load current in the primary-side elements
During the D’Ts interval when both output diodes conduct, inductor Lc stores energy (needed for ZVS to initiate the next DTs interval) and its current circulates around the primary-side elements—causing conduction loss

6. Diode switching analysis

7. Diode commutation: intervals 3 and 4

8. Waveforms: ZCS of D6

9. Intervals 3, 3', and 4'

10. Simplification of circuit model during 4'

11. Simplification of circuit model during 4'

12. State plane diagram of ringing during 4'

13. Waveforms: ZCS of D6

14. Approaches to snub the diode ringing (a) conventional diode snubber

15. Approaches to snub the diode ringing (b) conventional passive voltage-clamp snubber

16. Approaches to snub the diode ringing (c) simplify to one passive voltage-clamp snubber

17. Approaches to snub the diode ringing (d) improvement of efficiency in voltage-clamp snubber

18. Approaches to snub the diode ringing (e) active clamp lossless snubber

19. Approaches to snub the diode ringing (f) primary-side lossless voltage clamp

20. Another application of the ZVT: DC Transformer
Operate at a fixed conversion ratio with high duty cycle, leading to high efficiency—avoids the problems of circulating currents
Use other elements in the system to regulate voltage
5 V
1 V
PFC
350 V
DC-DC
Load
ZVT
AC line
Load
DC-DC
isolation
Load
DC-DC

21. Active clamp circuits
Can be viewed as a lossless voltage-clamp snubber that employs a current-bidirectional switch
See Vinciarelli patent (1982) for use in forward converter
Related to other half-bridge ZVS circuits
Can be added to the transistor in any PWM converter
Not only adds ZVS to forward converter, but also resets transformer better, leading to better transistor utilization than conventional reset circuit

22. The conventional forward converter
Max vds = 2Vg + ringing
Limited to D < 0.5
On-state transistor current is P/DVg
Magnetizing current must operate in DCM
Peak transistor voltage occurs during transformer reset
Could reset the transformer with less voltage if interval 3 were reduced

23. The active-clamp forward converter
Better transistor/transformer utilization
ZVS
Not limited to D < 0.5
Transistors are driven in usual half-bridge manner:

24. Approximate analysis: ignore resonant transitions, dead times, and resonant elements

25. Charge balance
Vb can be viewed as a flyback converter output. By use of a current-bidirectional switch, there is no DCM, and LM operates in CCM.

26. Peak transistor voltage
Max vds = Vg + Vb = Vg /D’
which is less than the conventional value of 2 Vg when D > 0.5
This can be used to considerable advantage in practical applications where there is a specified range of Vg

27. Design example 270 V ≤ Vg ≤ 350 V max Pload = P = 200 W
Compare designs using conventional 1:1 reset winding and using active clamp circuit

28. Conventional case Peak vds = 2Vg + ringing = 700 V + ringing
Let’s let max D = 0.5 (at Vg = 270 V), which is optimistic
Then min D (at Vg = 350 V) is (0.5)(270)/(350) =
The on-state transistor current, neglecting ripple, is given by  ig  = DnI = Did-on
with P = 200 W = Vg  ig  = DVg id-on
So id-on = P/DVg = (200W) / (0.5)(270 V) = 1.5 A

29. Active clamp case: scenario #1
Suppose we choose the same turns ratio as in the conventional design. Then the converter operates with the same range of duty cycles, and the on-state transistor current is the same. But the transistor voltage is equal to Vg / D’, and is reduced:
At Vg = 270 V: D = 0.5 peak vds = 540 V
At Vg = 350 V: D = peak vds = 570 V
which is considerably less than 700 V

30. Active clamp case: scenario #2
Suppose we operate at a higher duty cycle, say, D = 0.5 at Vg = 350 V. Then the transistor voltage is equal to Vg / D’, and is similar to the conventional design under worst-case conditions:
At Vg = 270 V: D = peak vds = 767 V
At Vg = 350 V: D = 0.5 peak vds = 700 V
But we can use a lower turns ratio that leads to lower reflected current in Q1:
id-on = P/DVg = (200W) / (0.5)(350 V) = 1.15 A