1. Data Converter

2. Design Of A Wireless Sensing

3. Analog to Digital (A/D) Converter
The analog to digital converter must take into account many factors, such as input signal, sampling rate, throughput, resolution, range, and gain. These will all be discussed shortly.
Input signal
Sampling rate
Throughput
Resolution
Range
Gain

4. Fundamentals of Sampled Data Systems
Analog-to-Digital converters (ADCs) translate analog quantities, wich are characteristic of most phenomen in the ‘’real world’’ to digital language, used in information processing, computing, data transmission, and control systems
Digital-to-Analog converters (DACs) are used in transforming transmitted or stored data, or the results of digital processing, back to ‘’real world’’ variables for control, information display, or further analog processing

5. Digital Number
Digital number used are all basically binary : that is, each ‘’bit’’ or unit of information
has one of two possible states.
These state are :
‘’off’’, ‘’false’’, or ‘’1’’
‘’on’’, ‘’true’’ , or ‘’0’’
It is also possible to represent the two logic state by two different levels of current ;
however, this is much less popular than using voltages .
Word are groups of levels representing digital numbers; the levels may appear
simultaneously in paralel , on a bus or groups of gate inputs or outputs,
serially (or in time sequence) on a single line,
as a sequence of parallel bytes (i.e. ‘’byte –serial’’) or nibbles (small bytes)
A unique parallel or serial grouping of digital levels, or a number, or code, is assigned
to each analog level which is quantized (i.e., represents a unique portion of the
analog range).

6. Typical Digital Code A typical digital code would be this array :
The meaning of the code, as either a number, a character, or a representation of an analog variable is unknow until the code and the conversion relationship have been defined

7. Unipolar Code

8. Bipolar Codes

9. Quantization: The Size of a Least Significant Bit (LSB)
Resolution N 2N
VOLTAGE
(10V FS)
ppm FS
% FS
dB FS
2-bit 4
2.5V 25
-12
4-bit 16
625mV
62.500
6.25
-24
6-bit 64
156mV
15.625
1.56
-36
8-bit
256
39.1mV
3.906
0.39
-48
10-bit
1.024
9.77mV (10mV)
977
0.098
-60
12-bit
4.096
2.44mV
244
0.024
-72
14-bit
16.384
610mV 61
0.0061
-84
16-bit
65.536
153mV 15
0.0015
-96
18-bit
38mV
0.0004
-108
20-bit
9.54mV (10mV) 1
0.001
-120
22-bit
2.38mV
0.24
-132
24-bit
596nV*
0.06
-144
The resolution of data converters

10. The Ideal Transfer Function (ADC)
The theoretical ideal transfer function for an ADC is a straight line, however, the practical ideal transfer function is a uniform staircase characteristic shown in Figure .

11. The Ideal Transfer Function (DAC)
The DAC theoretical ideal transfer function would also be a straight line with an infinite number of steps but practically it is a series of points that fall on the ideal straight line as shown in Figure

12. Sources of Static Error
Static errors, that is those errors that affect the accuracy of the converter when it is converting static (dc) signals, can be completely described by just four terms.
These are :
Each can be expressed in LSB units or sometimes as a percentage of the FSR
offset error,
gain error,
integral nonlinearity and
differential nonlinearity.

13. Offset Error - ADC
The offset error is efined as the difference between the nominal and actual offset points.

14. Offset Error - DAC
For a DAC it is the step value when the digital input is zero. This error affects all codes by the same amount and can usually be compensated for by a trimming process. If trimming is not possible, this error is referred to as the zero-scale error.

15. Gain Error - ADC
The gain error is defined as the difference between the nominal and actual gain points on the transfer function after the offset error has been corrected to zero. For an ADC, the gain point is the midstep value when the digital output is full scale,

16. Gain Error - DAC
For a DAC it is the step value when the digital input is full scale. This error represents a difference in the slope of the actual and ideal transfer functions This error can also usually be adjusted to zero by trimming.

17. Differential Nonlinearity (DNL) Error - ADC
DNL is the difference between an actual step width (for an ADC) and the ideal value of 1 LSB. Therefore if the step width is exactly 1 LSB, then the differential nonlinearity error is zero.
If the DNL exceeds 1 LSB  nonmonotonic (this means that the magnitude of the output gets smaller for an increase in the magnitude of the input)
If the DNL error of – 1 LSB there is also a possibility that there can be missing codes i.e., one or more of the possible 2n binary codes are never output.

18. Differential Nonlinearity (DNL) Error - DAC
The differential nonlinearity error shown in Figure is the difference between an actual step height (for a DAC) and the ideal value of 1 LSB. Therefore if the step height is exactly 1 LSB, then the differential nonlinearity error is zero

19. Integral Nonlinerity (INL) Error - ADC
The integral nonlinearity error shown in Figure is the deviation of the values on the actual transfer function from a straight line.
This straight line can be either a best straight line which is drawn so as to minimize these deviations or
it can be a line drawn between the end points of the transfer function once the gain and offset errors have been nullified (end-point linearity )

20. Integral Nonlinerity (INL) Error - DAC -
The name integral nonlinearity derives from the fact that the summation of the differential nonlinearities from the bottom up to a particular step, determines the value of the integral nonlinearity at that step.

21. Absolute Accuracy (Total) Error -ADC-
The absolute accuracy or total error of an ADC as shown in Figure is the maximum value of the difference between an analog value and the ideal midstep value.
It includes offset, gain, and integral linearity errors and also the quantization error in the case of an ADC

22. Absolute Accuracy (Total) Error -DAC-

23. Sampling Theory
Prior to the actual analog-to-digital conversion, the analog signal usually passes through some sort of signal conditioning circuitry which performs such functions as amplification, attenuation, and filtering.
The lowpass/bandpass filter is required to remove unwanted signals outside the bandwidth of interest and prevent aliasing.
There are two key concepts involved in the actual analog-to-digital and digital-to-analog conversion process:
An understanding of these concepts is vital to data converter applications.
discrete time sampling and
finite amplitude resolution due to quantization.

24. Sampling Theory
The system shown in Figure is real-time system ; i.e., the signal to the ADC is continuously sampled at a rate equal to fS, and the ADC presents a new sample to the DSP at this rate.
In order to maintain real-time operation, the DSP must perform all its required computation within the sampling interval, 1/fS, and present an output sample to the DAC before arrival of the next sample from the ADC.

25. The Need for a Sample-and-Hold Amplifier (SHA) Function
Most ADCs today have a built-in-sample-and-hold function, thereby allowing them to process ac signals.
This type of ADC is referred to as a sampling ADC
If the input signal to a SAR ADC (assuming no SHA function) changes by more than 1LSB during the conversion time (8ms is the example), the output data can have large errors, depending on the location of the code
Most ADC architectures are subject to this type of error – some more, some less – with the possible exception of flash converters having well-matched comparators

26. Input Frequency Limitations of Nonsampling ADC (Encoder)
This implies any input frequency greater than 9.7 Hz is subject to conversion errors, even though a sampling frequency of 100 kSPS is possible with the 8ms ADC (this allows an extra 2ms interval for an external SHA to reacquire the signal after coming out of hold mode).

27. Sample-and-Hold Function Required for Digitizing AC Signals
Sample-and-hold amplifier (SHA)
Track-and-hold amplifier (THA).

28. The Nyquist Criteria
A continuous analog signal is sampled at discrete intervals, fS,which must be carefully chosen to ensure an accurate representation of the original analog signal
The Nyquist criteria requiries that the sampling frequency be at least twice the highest frequency contained in the signal, or information about the signal will be lost
If the sampling frequency is less than twice the maximum analog signal frequency, a phenomen know as aliasing will occur
A signal with a maximum frequency .. must be sampled at a rate .... or information about the signal will be lost because of aliasing
Aliasing occurs whenever ...
A signal which has frequency components between .. and.... must be sampled at a rate in order to prevent alias components from overlapping the signal frequencies

29. Aliasing in Time Domain
In order to understand the implications of aliasing in both the time and frequency
domain, first consider the case of a time domain representation of a single tone sinewave sampled as shown in Figure

30. Matlab Example - 1

31. Matlab Example - 2

32. Matlab Example - 3

33. Matlab Example - 4

34. Matlab Example - 5

35. Aliasing in Frequency Domain
Consider the case of a single frequency sinewave of frequency fa sampled at
a frequency fs by an ideal impulse sampler.
Also assume that fs > 2fa as shown.
The frequency-domain output of the sampler shows aliases or images of the
original signal around every multiple of fs, i.e. at frequencies equal to |± Kfs ± fa|, K = 1, 2, 3, 4, .....

36. Baseband Antialiasing Filter
Baseband sampling implies that the signal to be sampled lies in the first Nyquist zone.
It is important to note that with no input filtering at the input of the ideal sampler, any frequency component (either signal or noise) that falls outside the Nyquist bandwidth in any Nyquist zone will be aliased back into the first Nyquist zone.
For this reason, an antialiasing filter is used in almost all sampling ADC applications to remove these unwanted signals.
The antialiasing filter transition band is therefore determined by the corner frequency fa, the stopband frequency fs – fa, and the desired stopband attenuation, DR. The required system dynamic range is chosen based on the requirement for signal fidelity.
For instance, a Butterworth filter gives 6-dB attenuation per octave for each
filter pole (as do all filters). Achieving 60 dB attenuation in a transition region between 1 MHz and 2 MHz (1 octave) requires a minimum of 10 poles—not a trivial filter, and definitely a design challenge.

37. Oversampling Relaxes Requirements on Baseband Antialiasing Filter
The effects of increasing the sampling frequency by a factor of K, while maintaining the same analog corner frequency, fa, and the same dynamic range, DR, requirement. The wider transition band (fa to Kfs – fa) makes this filter easier to design

38. Comparing a Nyquist rate (a) and Oversampling strategies (b)

39. Data Converter AC Error
The only errors (dc or ac) associated with an ideal N-bit data converter are those related to the sampling and quantization processes.
The maximum error an ideal converter makes when digitizing a signal is ±½ LSB.
The transfer function of an ideal N-bit ADC is shown in Figure

40. Quantization Noise as a Function of Time

41. FFT diagram of a multi-bit ADC with a sampling frequency FS
This noise is approximately Gaussian and spread more or less uniformly over the Nyquist bandwidth dc to fs/2.

42. Theoretical Signal-to-Quantization Noise Ratio of an Ideal N-Bit Converter

43. Procces Gain In many applications,
the actual signal of interest occupies a smaller bandwidth, BW.
If digital filtering is used to filter out noise components outside the bandwidth BW, then a correction factor (called process gain) must be included in the quation to account for the resulting increase in SNR.

45. SINAD, ENOB, SNR

46. Dynamic Range

47. Spurious Free Dynamic Range (SFDR)
Probably the most significant specification for an ADC used in a communications
application is its spurious free dynamic range (SFDR).
SFDR of an ADC is defined as the ratio of the rms signal amplitude to the rms value of the peak spurious spectral content measured over the bandwidth of interest.
SFDR is generally plotted as a function of signal amplitude and may be expressed
relative to the signal amplitude (dBc) or the ADC full-scale (dBFS) as shown in Figure

48. Aperture Time, Aperture Delay Time, and Aperture Jitter

49. Design a Low-Jitter Clock for High-Speed Data Converter
Many modern, high speed, high performance IC’s ADC’s require a low-phase-noise (low-jitter) clock that operates in the GHz range
Conventional crystal oscillators may provide a low jitter clock signal, but are not generally available in oscilating frequencies above 120 MHz
Typical high-speed data converter system

50. Jitter in clock signal degrades the ADC signal-to-noise ratio.
Jitter is generally defined as short-term, non-cumulative variation of the significant instant of a digital signal from its ideal position in time.
Figure illustrates a sampling clock signal that contains jitter. Jitter generated by the clock is caused by various internal noise sources, such as thermal noise, phase noise, and spurious noise.
A clock signal that has cycle-to-cycle variation in its duty cycle is said to exhibit jitter. Clock jitter causes an uncertainty in the precise sampling time, resulting in a reduction of dynamic performance.

51. How Clock Jitter Degrades ADC's Signal-to-Noise Ratio (SNR)

52. The functional diagram an integer-N PLL system
Consists of a phase detector (or comparator), an output charge-pump, a dual modulus prescalar, an N counter, and an R counter. The N counter consists of a main (M) counter and a swallow or auxiliary (A) counter. The N counter then works in conjunction with the dual modulus pre-scalar (P)

53. 1-Bit DAC: Changeover Switch (Single-Pole, Double Throw, SPDT)
Basic DAC Structures
1-Bit DAC: Changeover Switch (Single-Pole, Double Throw, SPDT)
switching an output between a reference and ground or between equal positive and negative reference voltages, as a 1-bit DAC
Such a simple device is a component of many more complex DAC structures, and is used, with oversampling, as the basic component in many of the sigma-delta DACs

54. The Comparator: A 1-Bit ADC
As a changeover switch is a 1-bit DAC, so a comparator is a 1-bit ADC.
If the input is above a threshold, the output has one logic value, below it has another.
Comparators used as building blocks in ADCs need good resolution which implies high gain. This can lead to uncontrolled oscillation when the differential input approaches zero. In order to prevent this, hysteresis is often added to comparators using a small amount of positive feedback

55. The Comparator: A 1-Bit ADC – cont.
Most modern comparators used in ADCs include a built-in latch which makes them sampling devices suitable for data converters.
A typical structure is shown in Figure
The latch thus performs a track-and-hold function, allowing short input signals to be detected and held for further processing.

56. Successive Aproximation ADCs
ADC Architectures
Flash Converters
Successive Aproximation ADCs
Pipelined ADCs
Integrating ADC
Sigma-Delta ADC

57. Classification ADC
Most ADC applications today can be classified into four broad market segments:
(a) data acquisition,
(b) precision industrial measurement,
(c) voiceband and audio, and
(d) “high speed” (implying sampling rates greater than about 5 MSPS).
A very large percentage of these applications can be filled by
A basic understanding of these, the three most popular ADC architectures—and their relationship to the market segments—is a useful supplement to the selection guides and search engines.
successive-approximation (SAR),
sigma-delta (-), and
pipelined ADCs

58. ADC Architectures, applications, resolution and sampling rates - 1

59. ADC Architectures, applications, resolution and sampling rates - 2

60. Flash Converters
Flash analog-to-digital converters, also known as parallel ADCS, are the fastest way to convert an analog signal to a digital signal.
An N-bit flash ADC consists of 2N resistors and 2N–1 comparators arranged as in Figure.
Since 2N–1 data outputs are not really practical, they are processed by a decoder to generate an N-bit binary output.
very large bandwidths.
consume a lot of power,
have relatively low resolution,
can be quite expensive

61. Architecture Detail
The reference voltage for each comparator is one least significant bit (LSB) greater than the reference voltage for the comparator immediately below it.

62. Sparkle Codes and Metastability
Normally, the comparator outputs will be a thermometer code, such as
Errors may cause an output like (i.e., there is a spurious zero in the result).
This out of sequence "0" is called a sparkle. This may be caused by imperfect input settling or comparator timing mismatch.
The magnitude of the error can be quite large.
Modern converters employ an input track-and-hold in front of the ADC along with an encoding technique that suppresses sparkle codes.
When a digital output of a comparator is ambiguous (neither a one nor a zero), the output is defined as metastable. Metastability can be reduced by allowing more time for regeneration. Gray-code encoding can also greatly improve metastability.

63. Successive-Approximation ADCs
The successive-approximation ADC is by far the most popular architecture for data-acquisition applications, especially when multiple channels require input multiplexing.
Modern IC SAR ADCs are available in resolutions from 8 bits to 18 bits, with sampling rates up to several MHz.
Output data is generally provided via a standard serial interface (I2C or SPI), but some devices are available with parallel outputs

64. Operation Algorithm
In order to process rapidly changing signals, SAR ADCs have an input sample-and-hold (SHA) to keep the signal constant during the conversion cycle.
The conversion starts with the internal D/A converter (DAC) set to midscale.
The comparator determines whether the SHA output is greater or less than the DAC output, and the result (the most-significant bit (MSB) of the conversion) is stored in the successive-approximation register (SAR) as a 1 or a 0.
The DAC is then set either to 1⁄4 scale or 3⁄4 scale (depending on the value of the MSB), and the comparator makes the decision for the second bit of the conversion
The result (1 or 0) is stored in the register, and the process continues until all of the bit values have been determined.
At the end of the conversion process, a logic signal (EOC, DRDY, BUSY, etc.) is asserted.
The acronym, SAR, which actually stands for successive-approximation register—the logic block that controls the conversion process—is universally understood as an abbreviated name for the entire architecture.

65. Basic Successive-Approximation ADC
The overall accuracy and linearity of the SAR ADC are determined primarily by the internal DAC’s characteristics

66. Functional block Diagram of a modern 1-MSPS SAR
The sequencer allows automatic conversion of the selected channels, or channels can be addressed individually if desired. Data is transferred via the serial port. SAR ADCs are popular in multichannel data-acquisition applications

67. Pipelined ADCs for High-Speed Applications (Sampling Rates Greater than 5 MSPS)
The low-power CMOS pipelined converter is the ADC of choice, not only for the video market but for many others as well
Today, markets that require “high speed” ADCs include many types of:
The pipelined ADC has its origins in the subranging architecture
A block diagram of a simple 6-bit, two-stage subranging ADC is shown in Figure
instrumentation applications (digital oscilloscopes, spectrum analyzers, and medical imaging).
video, radar, communications (IF sampling, software radio, base stations, set-top boxes, etc.),
consumer electronics (digital cameras, display electronics, DVD, enhanced-definition TV, and high-definition TV)

68. 6-bit, two-stage subranging ADC
The output of the SHA is digitized by the first-stage 3-bit sub-ADC (SADC)—usually a flash converter.
The coarse 3-bit MSB conversion is converted back to an analog signal using a 3-bit sub-DAC (SDAC).
Then the SDAC output is subtracted from the SHA output, the difference is amplified, and this “residue signal” is digitized by a second-stage 3-bit SADC to generate the three LSBs of the total 6-bit output word

69. Residue waveform at input of second-stage SADC
This waveform is typical for a low-frequency ramp signal applied to the analog input of the ADC.
In order for there to be no missing codes, the residue waveform must not exceed the input range of the second-stage ADC, (Figure A).
The situation shown in Figure B will result in missing codes when the residue waveform goes outside the range of the N2 SADC, “R,” and falls within the “X” or “Y” regions—which might be caused by a nonlinear N1 SADC or a mismatch of interstage gain and/or offset.

70. The error-corrected subranging ADC architecture
A basic 6-bit subranging ADC with error correction is shown in Figure, with the second-stage resolution increased to 4 bits, rather than the original 3 bits. Additional logic, required to modify the results of the N1 SADC when the residue waveform falls in the “X” or “Y” overrange regions, is implemented with a simple adder in conjunction with a dc offset voltage added to the residue waveform. In this arrangement, the MSB of the second-stage SADC controls whether the MSBs are incremented by 001 or passed through unmodified.

71. “Pipelined” architecture
In order to increase the speed of the basic subranging ADC, the “pipelined” architecture has become very popular.
This pipelined ADC has a digitally corrected subranging architecture — in which each of the two stages operates on the data for one-half of the conversion cycle, and then passes its residue output to the next stage in the “pipeline” prior to the next phase of the sampling clock.
The interstage track-and-hold (T/H) serves as an analog delay line — it is timed to enter the hold mode when the first-stage conversion is complete. This allows more settling time for the internal SADCs, SDACs, and amplifiers, and allows the pipelined converter to operate at a much higher overall sampling rate than a nonpipelined version.

72. Generalized pipeline stages and timing

73. Clock Issues in Pipelined ADCs
Notice that the phases of the clocks to the T/H amplifiers are alternated from stage to stage such that when a particular T/H in the ADC enters the hold mode it holds the sample from the preceding T/H, and the preceding T/H returns to the track mode. The held analog signal is passed along from stage to stage until it reaches the final stage in the pipelined ADC

74. Dual Slope ADCs
The dual-slope ADC architecture was truly a breakthrough in ADCs for high resolution applications such as digital voltmeters, etc.
The input signal is applied to an integrator; at the same time a counter is started, counting clock pulses. After a pre-determined amount of time (T), a reference voltage having opposite polarity is applied to the integrator. At that instant, the accumulated charge on the integrating capacitor is proportional to the average value of the input over the interval T.

75. Dual Slope ADCs – cont.
The integral of the reference is an opposite-going ramp having a slope of VREF/RC. At the same time, the counter is again counting from zero. When the integrator output reaches zero, the count is stopped, and the analog circuitry is reset. Since the charge gained is proportional to VIN · T, and the equal amount of charge lost is proportional to
VREF · tx, then the number of counts relative to the full scale count is proportional to tx/T, or VIN/VREF. If the output of the counter is a binary number, it will therefore be a binary representation of the input voltage.

76. - ADC architecture
Modern - ADCs for applications requiring high resolution (16 bits to 24 bits) and effective sampling rates up to a few hundred hertz.
High resolution, together with on-chip programmable-gain amplifiers (PGAs), allows the small output voltages of sensors — such as weigh scales and thermocouples — to be digitized directly.
Proper selection of sampling rate and digital filter bandwidth also yields excellent rejection of 50-Hz and 60-Hz power-line frequencies.
- ADCs offer an attractive alternative to traditional approaches using an instrumentation amplifier (in-amp) and a SAR ADC.

77. The basic concepts - ADC architecture - 1
Figure A shows a noise spectrum for traditional “Nyquist” operation, where the ADC input signal falls between dc and fS/2, and the quantization noise is uniformly spread over the same bandwidth

78. The basic concepts - ADC architecture - 2
In Figure B, the sampling frequency has been increased by a factor, K, (the oversampling ratio), but the input signal bandwidth is unchanged.
The quantization noise falling outside the signal bandwidth is then removed with a digital filter.
The output data rate can now be reduced (decimated) back to the original sampling rate, fS. This process of oversampling, followed by digital filtering and decimation, increases the SNR within the Nyquist bandwidth (dc to fS/2).
For each doubling of K, the SNR within the dc-to-fS/2 bandwidth increases by 3 dB.

79. The basic concepts - ADC architecture - 3
Figure C shows the basic - architecture, where the traditional ADC is replaced by a - modulator.
The effect of the modulator is to shape the quantization noise so that most of it occurs outside the bandwidth of interest, thereby greatly increasing the SNR in the dc-to-fS/2 region.

80. First-order sigma-delta ADC
The heart of this basic modulator is a 1-bit ADC (comparator) and a 1-bit DAC (switch).
The output of the modulator is a 1-bit stream of data.
The noise-shaping function by acting as a low-pass filter for the signal and a high-pass filter for the quantization noise.

81. Sigma-Delta Modulator Waveforms
Because of negative feedback around the integrator, the average value of the signal at B must equal VIN. If VIN is zero (i.e., midscale), there are an equal number of 1s and 0s in the output data stream. As the input signal goes more positive, the number of 1s increases, and the number of 0s decreases. Likewise, as the input signal goes more negative, the number of 1s decreases, and the number of 0s increases. The ratio of the 1s in the output stream to the total number of samples in the same interval—the ones density—must therefore be proportional to the dc value of the input

82. Example:
Analog input 3/8

83. Second-order - modulator

84. 24-bit - Converter

85. Some General Trends in Data Converters
The general trends in data converters are summarized in Figure :

86. Low Power, Sleep, and Standby Modes
In order to conserve power, especially in battery-powered applications, most modern data converters have some type of low-power, sleep, or standby mode, where the major portion of the internal circuitry is powered down—usually initiated by
the application of a signal to one of the pins,
software control via internal control registers.
additional power savings can be achieved by disabling some or all of the external clocks.
Sleep-mode power supply current  from a few μA to tens of mA depending upon the normal-mode power dissipation.
Recovery time from the sleep mode, or power-up time  but generally is in the order of a few μs to 100 μs.

87. ADC Serial Output Interfaces
Serial outputs on SAR-based and Σ-Δ ADCs since their conversion architecture is essentially serial.
If an ADC is operating continuously, the period of the sampling clock must be long enough to transfer all the serial data across the interface at the interface data rate, with some appropriate amount of headroom.
Example:
A 16-bit, 1-MSPS sampling ADC requires a serial output data rate of at least 16 MHz, which would not be a problem with most modern P, Cor DSPs.

88. ADC Parallel Output Interfaces
Parallel ADC output interfaces are popular, straightforward, and must be used when the product of sampling rate and resolution exceeds the capacity available serial links.
Example:
Using a maximum LVDS serial data link of 600 Mbits/s requires parallel data transmission for resolutions/sampling rates greater than 8 bits at 75 MSPS, 10 bits at 60 MSPS, 12 bits at 50 MSPS, 14 bits at 43 MSPS, 16 bits at 38 MSPS, etc.

89. Data Converter Voltage References
The accuracy of a data converter is determined by a voltage reference of some sort.
An exception to this, of course, is an ADC which operates in a ratiometric mode, where both the input signal and input range scale proportionally to the reference.
Example:
Voltage references have a major impact on the performance and accuracy of analog systems. A ±5-mV tolerance on a 5-V reference corresponds to ±0.1% absolute accuracy—only 10 bits. For a 12-bit system, choosing a reference that has a ±1-mV tolerance may be far more cost effective than performing manual calibration, while both high initial accuracy and calibration will be necessary in a system making absolute 16-bit measurements.

90. Ratiometric
ADC can be driven from a single supply voltage which is also used to excite the remote bridge. Both the analog input and the reference input to the ADC are high impedance and fully differential. By using the + and – SENSE outputs from the bridge as the differential reference to the ADC, the reference voltage is proportional to the excitation voltage which is also proportional to the bridge output voltage.

91. Some Popular ADC/DAC Reference Options

92. converter which requires an external reference
converter which requires an external reference. It is generally recommended that a suitable decoupling capacitor be added close to the ADC/DAC REF IN pin
converter that has an internal reference, where the reference is also brought out to a pin on the device. This allows it to be used other places in the circuit,
provided the loading does not exceed the rated value.
converter which can use either the internal reference or an external
one, but an extra package pin is required. If the internal reference is used, REF OUT is simply externally connected to REF IN, and decoupled if required.
If an external reference is used as shown, REF OUT is left floating, and the external reference decoupled and applied to the REF IN pin.
shows an arrangement whereby an external reference can override the
internal reference using a single package pin. The value of the resistor, R, is typically a few kΩ, thereby allowing the low impedance external reference to override the internal one when connected to the REF OUT/IN pin.
shows how the external reference is connected to override the internal reference.

93. Types of Voltage References
Basic Bandgap Reference
Simple Diode Reference Circuits

94. Selecting an A/D Converter
The selection checklist can be broken up into two areas —
primary facts which cannot be compromised, and
secondary factors which may allow the designer some flexibility
Primary
• What is the required level of system accuracy?
• How many bits of resolution are required?
• What is the nature of the analog input signal?
• How fast must the converter operate (conversion speed)?
• What are the environmental conditions?
• Is a track-and-hold circuit required?

95. Selecting an A/D Converter
Secondary
• Does the system have multiple channels?
• Should the reference be internal or external?
• What are the drive amplifier requirements?
• What are the digital interface requirements?
• What type of digital output format is required?
• What are the timing conditions?

96. Caracteristics ADC

97. How to Save Power ?
The serial interface consists of the CS, SCLK, and SDATA lines
A normal conversion requires sixteen serial clock pulses for completion.
shows how the power-down mode can be entered by controlling the CS signal

98. Texas Instruments - ADS7807

99. Analog Devices - AD7466
The AD7466, a micropower, 12-bit SAR-type ADC housed in a 6-lead SOT-23 package. It can be operated from 1.6 V to 3.6 V and is capable of throughput rates of up to 200 kSPS.
The current consumption in power-down mode is typically 8 nA. The AD7466 consumes 0.9 mW max when operating at 3 V, and 0.3 mW max for 1.8 V operation at 100 kSPS.