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Page 1 27 th February 2007 On-Board Calibration System for the Range Delay of the BepiColombo KaT G.Boscagli (ESA-ESTEC) M.Mascarello (AAS-I) 27 th February.

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Presentation on theme: "Page 1 27 th February 2007 On-Board Calibration System for the Range Delay of the BepiColombo KaT G.Boscagli (ESA-ESTEC) M.Mascarello (AAS-I) 27 th February."— Presentation transcript:

1 Page 1 27 th February 2007 On-Board Calibration System for the Range Delay of the BepiColombo KaT G.Boscagli (ESA-ESTEC) M.Mascarello (AAS-I) 27 th February 2007 MORE Team Meeting

2 Page 2 27 th February 2007 Introduction for BepiColombo KaT On-board Calibration (for Ranging Delay)

3 Page 3 27 th February 2007 Approach for On-Board Calibration  The approach hereafter preliminary analysed is based on the idea of implementing this function directly inside the KaT unit: Target  Calibration as KaT Internal Unit Function  The idea is to include inside the KaT unit both the SSPA and the Diplexer function. In general more compact solutions improve S/C design (mass, interfaces routing, etc); it is believed that also calibration performances should be improved following this approach. NOTE – In this way the calibration doesn’t take into account the wave-guides (from-to-antenna) and the antenna itself. They are outside the calibration loop. Which is the contribution of wave-guides and antenna in the overall end-to-end ranging budget error? At present it is understood that the main effect to be considered is related to the input/output mismatching variation due to temperature variation. This might cause multi-path effects and error in the end-to-end ranging measurements.

4 Page 4 27 th February 2007 Approach for On-Board Calibration  For this reason (Calibration as an internal KaT function) the KaT unit must be commanded (by the on-board computer) in two different modes:  Nominal Mode: RF link via antenna, unit in coherent mode (down-link coherent with the uplink both for carrier and ranging signal) when RX in Tracking Mode.  Calibration Mode: RX and TX in Loop-back Configuration, unit running with the internal oscillator, RX coherent (in tracking) with the loop-back signal from the TX  The approach hereafter proposed is based on the use of PN regenerative ranging, the reasons are:  The use of this ranging scheme simplifies the calibration scheme in particular for the ambiguity resolution (when compared with other approaches as the ESA STD or the NASA Tone Ranging). Note - The ambiguity must be solved since the KaT loop-back delay (TX-RX) is expected of the order of microseconds, while the WBRS band should be in the range MHz. This comment might not be valid anymore in case the group delay variation (versus environmental conditions and including aging) is inside the WBRS ambiguity resolution. This is difficult to be predicted at this stage.  The PN regenerative ranging is already implemented inside the BepiColombo X/X/Ka Deep Space Transponder, so it can be easily re-used for the KaT unit simplifying the multi-frequency operation as needed by BepiColombo for plasma cancellation.

5 Page 5 27 th February 2007 Impact of Calibration on BepiColombo KaT Front-End Architecture NOTE: this section has been written without considering the current subsystem architecture

6 Page 6 27 th February 2007 Impact of Calibration on KaT Front-End Architecture RF Budget TX2W33.01dBm TX coupler30dB3.01dBm Attenuator- T130dB-26.99dBm Passive Mixer10dB-36.99dBm Attenuator- T260dB-96.99dBm RX coupler50dB dBm Preliminary RF Power Budget (TX Filter neglected) T2 is indicated as variable attenuator, it might be commanded for selecting the proper RF power (at RX input) for calibration: minimum value around -146 dBm (nominal value around -125 dBm, TBC). 34 GHz Amplifier TX Filter T1 X T2 To Ant 2 GHz 32 GHz RX Filter 2 Watt SSPA LNA TX coupler RX coupler Amplifier Gain Control (TBC) Loop-Back Path

7 Page 7 27 th February 2007 Impact of Calibration on KaT Front-End Architecture  According to the proposed approach, the new circuits/functions to be developed are:  The functions in the light-blue boxes: 2 GHz LO, T1, T2, Mixer  The functions in the light-green boxes: TX coupler, RX coupler NOTE - The functions indicated in the light-red boxes represent the Diplexer and they are present in any case. In the current subsystem baseline (see dedicated slides by AAS-I on this issue), the Diplexer is indicated as external to the KaT; the advantage of having it integrated inside the KaT unit is clear from a calibration point of view and for a more compact solution.  According to the architecture as proposed in the previous slide, when in calibration mode the loop-back signal is routed back from the TX to the RX side and to the antenna as well. NOTE – It might be useful (TBC) to introduce the control of the TX Power (2 Watt SSPA) to minimise the TX power via antenna when in calibration mode; question: is the SSPA delay dependant on the selected Gain/Output-Power?

8 Page 8 27 th February 2007 Impact of Calibration on KaT Frequency Plan  The KaT frequency plan must be carefully studied in order to simplify the generation of the 2 GHz LO signal and to avoid internal RFI issues. NOTE - For instance considering the Cassini KaT frequency plan (next slide) we observe that the 1 st IF chain is almost at the same frequency of the LO signal for calibration.  The Cassini KaT turn-around ratio (next slide) is not included in the current CCSDS/ECSS recommendations for TT&C applications. The values from the current ECSS-E-50-05B (Radio Frequency and Modulation, draft issue under public review) are:  The Ka/Ka turn-around ratio values are under discussion also in the frame of CCSDS, at present the draft recommendation (January 2007) is to use 3599/3344 and 3599/3360

9 Page 9 27 th February 2007 Cassini KaT Frequency Plan

10 Page th February 2007 CURRENT COMMUNICATIONS SUBSYSTEM BASELINE (from BepiColombo SRR Data Package) This section has been provided by AAS-I (Marco Mascarello) Question: Are there any difficulties (due to on-board baseline architecture) for implementing the above proposed approach for calibration?

11 Page th February 2007 Communications Subsystem Current baseline

12 Page th February 2007 Communications Subsystem (Option KaT Amplifier)

13 Page th February 2007 KaT on board calibration including Triplexer Including a Triplexer inside the KaT, it would be possible to calibrate all the paths till the antenna interface. 34 GHz Amplifier TX Filter T1 X T2 To Ant 2 GHz 32 GHz RX Filter 2 Watt SSPA LNA circulator coupler Ka TRIPLEXER KaT assembly From Ka-Band TWTA (X/X/Ka DST)

14 Page th February 2007 Triplexer (from current BepiColombo SRR Data Package)  The Ka-Band Triplexer is a 4 port device in charge of splitting input and output signals. It will consist of a new development for BepiColombo based on existing technology.  The splitting will be accomplished by an E-plane trifurcation. Each sub-band will be selected by an H plane filter. The foreseen useful bandwidth of the filters will be the following:  Rx Filter : 50 MHz (TBC) within to MHz  Tx1 Filter : 200 MHz (TBC) within to MHz  Tx2 Filter : 50 MHz within (TBC) to MHz  As for the X-Band diplexer, the mechanical concept will be two symmetrical pieces. Interfaces will be standard WR28 waveguide flanges. The estimated dimensions for the assembly are 75 x 45 x 23.1 mm, while the estimated maximum mass should be less than 65 g.

15 Page th February 2007 Concern  The above solution based on the Triplxer inside the KaT unit shows an important drawback: The Ka-band DST signal is applied to the Antenna through the KaT unit. This represents a blocking point !  Other solution must be addressed, for instance: 1.Keeping the Triplexer external to the KaT unit (calibration not anymore an internal KaT function). 2.Analysing different mixing approach between DST and KaT signals at HGA input.

16 Page th February 2007 BepiColombo X/X/Ka DST PN Regenerative Ranging  1999 JPL  Balanced Weighted-Voting Tausworthe (v=2 and 4)

17 Page th February 2007 Introduction to Pseudo Noise (PN) Ranging Sequence The term “Pseudo-Noise (PN) ranging” refers in a strict sense to the use of a ranging-sequence system in which the ranging sequence is a logical combination of the so-called range clock-sequence and several Pseudo- Noise (PN) sequences. The range clock sequence is the alternating +1 and –1 sequence of period 2 chips. A Pseudo-Noise (PN) sequence is a binary  1 sequence of period L whose periodic autocorrelation function has peak value +L and all (L–1) off-peak values equal to –1. Range Clock Frequency =

18 Page th February 2007 Example for the introduction of the Titsworth/Tausworthe generation scheme As an example, considering the following component sequences of period 2, 3 and 5, respectively (the first period of each sequence is underlined): Combined by majority logic give the following period-30 sequence: Component Sequences or Probe Sequences PN Sequence Seq. Gen. # 1 Seq. Gen. # 2 Seq. Gen. # 3 Introduction to Pseudo Noise (PN) Ranging Sequence

19 Page th February 2007  Note that the period T of the PN sequence obtained with the Tausworthe scheme is given by: with LCM = Least Common Multiple 30 in the above example  The correlation of this sequence (considered as +/-1 sequence) with the component/probe sequences gives the following results:  Note that = 10 operations of correlation are required instead of the 30 operations needed in the “classical” approach. In fact, only 9 decisions are required because of the antipodal result of the sequence of period-2 (the clock sequence). Only one of the two operation of correlation must be performed because the other correlation will be the negative of the other. Importance of having prime length component sequences Introduction to Pseudo Noise (PN) Ranging Sequence Example for the introduction of the Titsworth/Tausworthe generation scheme

20 Page th February 2007  The ranging sequence is acquired by the receiver as the result of correlations between the received sequence and certain ±1 periodic sequences (and their cyclic shifts) whose periods are divisors of the ranging  sequence period and that we will refer to as probing sequences.  The probing sequences are related in some manner to the ranging sequence, e.g., the ranging sequence might be the sequence resulting from some sort of voting by the chips of all the probing sequences at the same chip time.  The probing sequences must have the property that when all these “in-phase” decisions are correctly made, then these decisions determine the delay (modulo the ranging sequence period L) in chips of the received ranging sequence relative to the corresponding model of the ranging sequence. The (one-way) ambiguity (U) due to the period of the ranging sequence in meters is For example, with L = 1,009,470 chips and Hz, U  75,710,000 m or about 75,710 km. Introduction to Pseudo Noise (PN) Ranging Sequence = ranging clock frequency chip rate c = speed of the light More in general we can state that:

21 Page th February 2007 The 1999 JPL PN Ranging scheme (Tausworthe scheme)  The combining logic is based on the following rule: the ranging-sequence chip is a +1 if and only if either C1 has a +1 at that position or all five of the sequences C2, C3, C4, C5 and C6 have a +1 at that position, or both. Titsworth/Tausworthe generation scheme  C1, C2, … C6 are the so called Probing Sequences. In literature this sequence can be indicated also as JPL 99 or Taus

22 Page th February 2007 The 1999 JPL PN Ranging scheme (Tausworthe scheme)  It is obvious from this combinational rule that the range clock will be strongly correlated with the ranging sequence, which facilitates locking on to the range clock at the receiver.  Since the component sequences C2, C3, C4, C5 and C6 are all PN sequences with relatively prime periods 7, 11, 15, 19 and 23, respectively, the period of the 1999 JPL ranging sequence is L =2x7x11x15x19x23 = 1,009,470 chips.  The probing sequences in the 1999 JPL PN ranging-scheme are the range clock sequence together with the five component PN sequences.  The total number of correlation operations required for the probing sequences, excluding the range clock, is thus = 75.

23 Page th February 2007 The 1999 JPL PN Ranging scheme (Tausworthe scheme) Correlation characteristics and spectrally relevant properties of the ranging sequence and probing sequences The spectrum shows a powerful clock component at half the chip rate and below a noisy floor originating from the combination process with the other probing sequences. The fact the range clock is strongly correlated with the ranging sequence will facilitate locking on to the range clock at the receiver. The chip is square-wave shaped. Residual carrier Mod index = 0.82 rad-pk Clock Components at ±f RC Chip Rate at ± f Chip_Rate = 2.5 Mcps

24 Page th February 2007 Weighted-Voting Tausworthe PN Ranging-Sequence Scheme The Weighted-Voting Tausworthe sequences are derived from the 1999 JPL PN Ranging sequence with an apparently small modification on the vote logic.  The selection of different value for the clock vote (v=2 or 4) provides:  flexibility in the choice of the strength of the range-clock component in the ranging sequence  different level for the power allocated to the clock and the other ranging spectral components. v

25 Page th February 2007 Balanced Weighted-Voting Tausworthe PN Ranging-Sequence Scheme The Balanced Weighted-Voting Tausworthe sequences are derived from the Weighted-Voting Tausworthe sequences (scheme above) with an apparently small modification on the polarity of some probe sequences.  As the 1999 JPL PN Ranging scheme (Tausworthe scheme) also the Weighted-Voting Tausworthe PN Ranging-Sequence Schemes (both for v=2 and 4) present a DC component.  A simple way to reduce the imbalance in the ranging sequence (and to produce what we call the Balanced Weighted-Voting Tausworthe ranging-sequence scheme) is choosing the PN probing sequences with the following first periods: C1 = +1  1 C2 =  1  1 +1  1 -C3 =  1  1   1 +1  1  C4 =  1  1  1    1  1 +1  1 +1 C5 =  1 +1  1 +1  1  1  1    1  1 -C6 =  1  1  1  1  1 +1  1 +1  1   1   1 +1  Note - The key to elimination of imbalance is the fact the negative of a real sequence has the same autocorrelation function as the original sequence.

26 Page th February 2007 BepiColombo X/X/Ka DST: Code Phase Acquisition  the Chip Tracking Loop (CTL) for ranging code clock component phase and frequency recovery  the In-phase Integrator output is provided to Code Correlators: Six Correlators running in parallel for probe sequences (C1,…. C6) position recovery  the Down-link Code Generator (In this case only the JPL99 case is represented) The current model of BepiColombo X/X/Ka DST is programmable and can handle the different schemes: JPL99, BT2 and BT4. The Regenerative Ranging Channel is composed by: From Carrier Quadrature branch

27 Page th February 2007 BepiColombo X/X/Ka DST: Chip Tracking Loop (CTL) The mid-phase integrator output is multiplied by +/-1 in order to provide the right correction to the loop. In a certain way the multiplication by +/-1 replaces the transition detector typical of a DDTL, considering that the PN sequence resembles a square-wave. CTL NCO Base Frequency Filtered Loop Error Scaled Carrier Loop Error Quadrature Carrier Branch Output

28 Page th February 2007 BepiColombo KaT Calibration based on PN Regenerative Ranging

29 Page th February 2007 Impact of Calibration on BepiColombo KaT Baseband Processing  We need a separate PN code generator on the TX side clocked by the on- board oscillator  In the current X/X/Ka DST design the TX PN code is generated coherently with the received up-link PN code (see previous slide).  The TX PN NCO and the RX PN NCO must be clocked with the same oscillator, avoiding any timing error between the two signals.  At the start of the calibration procedure (defined by a strobe signal common to RX and TX processing functions) the two PN code generators (RX and TX) must be identically initialised.  The loop back ranging signal acquired by the RX provides the delay from TX to RX (Loop-Back Delay).  The PN code phase acquisition (using the Probe Sequences) is used for ambiguity resolution  The phase difference between TX and RX ranging clock provides the accurate delay measurement

30 Page th February 2007 Impact of Calibration on BepiColombo KaT Baseband Processing  The phase difference between the RX and TX PN Ranging Clock can be measured using the filtered phase error loop term of the CTL CTL Detector RX NCO 1/S X X + + B (nominal chip rate) α β TX NCO On-Board Clock To RX PN Code Generator To TX PN Code Generator B (nominal chip rate) KdKd K NCO CTL second order loop E

31 Page th February 2007 Impact of Calibration on BepiColombo KaT Baseband Processing  Open Loop CTL Transfer Function  In the X/X/Ka DST the CTL is digitally implemented inside the RX Digital Section (T s is the loop sampling time), using the Z transfer function we have CTL Detector RX NCO N-BIT X X + + B (nominal chip rate) αTs βTs2βTs2 F CLK =1/T s TsTs KdKd E CTL second order loop: digital representation

32 Page th February 2007 Impact of Calibration on BepiColombo KaT Baseband Processing  After the transient phase, the error term E provides the measurement of the delay between the TX and RX ranging clock signal. In radiant we can write:  While in time we have:  It is evident that for typical loop sampling time of the order of 40 MHz and N=32 bit NCO the phase/time resolution is well below the required BepiColombo ranging delay accuracy. The calibration resolution in time due to the digital loop implementation is:  The Probe Sequence acquisition phase and the CTL error term (E) must be transmitted via telemetry down-link (using the X/X/Ka DST link). This information (after proper post-processing) can be used (on-ground) to evaluate the accurate Loop-Back delay.

33 Page th February 2007 Impact of Calibration on BepiColombo KaT Baseband Processing  Notice that the KaT Ranging Delay (RX => TX) and the Loop Back Ranging Delay (TX => RX) might be different, this is due to: 1.The TX/RX different paths (between nominal and calibration mode) in the Front-End (Attenuators, Mixer, Couplers) 2.Different routing of the signal in the baseband digital processing (ASIC gates).  Probably the first contribution could be kept small (and negligible also under variations of environmental conditions) in terms of overall error budget. This to avoid further complications in terms of calibration.  Also the second contribution (inside the DSP) might be kept negligible; however if not negligible, the delta (between the KaT Ranging Delay and the Loop Back Ranging Delay) can be measured at unit level in the LAB. Notice that this contribution is almost independent from the temperature since it is related to the clock drift (Note - the X/X/Ka DST is embarking an OCXO).

34 Page th February 2007 CONCLUSIONS

35 Page th February 2007  In order to improve the calibration performances and to minimise the on-board complexity it is suggested to integrate inside the KaT unit the SSPA, the Diplexer (*), the RF Calibration Front-End (Attenuators, Couplers, mixer, LO). (*) The possibility to integrate the Diplexer/Triplexer is not clear (under discussion).l  The use of PN Ranging (as per X/X/Ka DST) simplifies the calibration function in particular for ambiguity resolution.  Minor changes are foreseen for the base-band digital signal processing(*). The approach is to transmit the CTL error term via TLM link for post-processing at the G/S. (*) However these have an impact on the current X/X/Ka DST FPGA/ASIC Conclusions


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