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Electric Drives1 ELECTRIC DRIVES Ion Boldea S.A.Nasar 1998

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Electric Drives2 PM AND RELUCTANCE SYNCHRONOUS MOTOR DRIVES 11.1. INTRODUCTION PM and reluctance synchronous motors are associated with PWM voltage - source inverters in variable speed drives. While PM - SM drives enjoy by now world - wide markets, reluctance synchronous motor (RSM) drives still have a small share of the market in low power region, though recently high saliency (and performance) RSMs with q axis PMs have been successfully introduced for wide constant power speed range applications (such as spindle drives). High performance applications are in general provided with PM - SM drives. RSM drives may be used for general applications as the costs of the latter are lower than of the former and, in general, close to the costs of similar IM drives. In what follows we will deal first with PM - SM drives and after that with RSM drives. Constant power operation will be discussed at the end of the chapter for both types of synchronous motors.

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Electric Drives3 11.2. PM - SM DRIVES: CLASSIFICATIONS Basically we may distinguish three ways to classify PM - SM drives: with respect to current waveform, voltage - frequency correlation, motion sensor presence. From the point of view of current waveform we distinguish: rectangular current control - figure 11.1.a - the so called brushless d.c. motor drive - q = 1 slot per pole per phase, surface PM rotor; sinusoidal current control - figure 11.1.b - the so called brushless a.c. drive - q 2 slots per pole per phase. From the point of view of motion sensor presence there are: drives with motion sensors; drives without motion sensors (sensorless). Finally sinusoidal current (brushless a.c.) drives may have: scalar (V / f) control - a damper cage on the rotor is required; vector control (current or current and voltage); direct torque and flux control (DTFC). The stator current waveforms - rectangular or sinusoidal (figure 11.1) - have to be synchronized with the rotor position.

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Electric Drives4 Figure 11.1. Rectangular or sinusoidal current control; a - advance angle

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Electric Drives5 Scalar control (V / f) is related to sinusoidal current control without motion sensors (sensorless) (figure 11.2). Figure 11.2. V / f (scalar) control for PM - SM (and for RSM) with torque angle increment compensation

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Electric Drives6 For faster dynamics applications vector control is used (figure 11.3). Figure 11.3. Basic vector control of PM - SM (or for RSM) 1 - with motion sensor2 - sensorless

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Electric Drives7 Figure 11.4. Direct torque and flux control (DTFC) of PM - SMs (and RSMs)

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Electric Drives8 To simplify the motor control, the direct torque and flux control (DTFC for IMs) has been extended to PM - SM (and to RSMs) as torque vector control (TVC) in [10]. The stator flux and torque direct control leads to a table of voltage switchings (voltage vector sequence). Vector rotation has been dropped but flux and torque observers are required. While speed is to be observed, rotor position estimation is not required in sensorless driving. Again, fast flux and torque control may be obtained even in sensorless driving. Rectangular current control and sinusoidal current control (through vector control or DTFC) are going to be detailed in what follows for motion sensor and sensorless driving.

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Electric Drives9 11.3. RECTANGULAR CURRENT CONTROL (BRUSHLESS D.C. MOTOR DRIVES) Rectangular current control is applied to nonsinusoidal (trapezoidal) PM - e.m.f. waveform - typical to concentrated coil stator windings (three slots per pole: q = 1). To reduce torque pulsations rectangular current is needed. With rectangular current, however, the reluctance torque production is not so efficient and thus nonsalient pole (surface PM) rotors is preferred. The motor phase inductance L a is [1]: (11.1) (11.2) where: W 1 - turns per phase, L - stack length, - pole pitch, g - airgap, K c - Carter coefficient, h PM - PM radial thickness, p - pole pairs, L sl - leakage inductance. The mutual inductance between phases, L ab, is: (11.3) Thus the cycling inductance L s (per phase) for two phase conduction is: (11.4)

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Electric Drives10 11.3.1. Ideal brushless d.c. motor waveforms In principle the surface PM extends over an angle PM less than 180 0 (which represents the pole pitch, figure 11.1). The two limits of PM are 2 /3 and. Let us suppose that the PM produces a rectangular airgap flux distribution over PM = (180 0 ) (figure 11.5.a). The stator phase m.m.f. is supposed to be rectangular, a case corresponding to q = 1 (three slots per pole). Consequently the PM flux linkage in the stator winding PM ( er ) varies linearly with rotor position (figure 11.5.b). Finally the phase e.m.f. E a is rectangular with respect to rotor position (figure 11.5.c). For zero advance angle a = 0 (figure 11.1) - the phase currents are in phase with the e.m.f. (i a, E a ) for motoring (figure 11.5.d). At any instant, due to assumed instantaneous current commutation, only two phases are in conduction. Consequently only two thirds of the PMs are utilized. So the back core flux is 50% larger than necessary. On the other hand, if the currents would extend over 180 0 (with three phases conducting at any time) for 120 0 wide PMs, it would mean 50% more than necessary copper losses for a given torque.

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Electric Drives11 Figure 11.5. Ideal waveforms for BLDC (brushless d.c. motor) a.) PM airgap flux densityb.) PM flux per phase ac.) e.m.f. in phase ad.) ideal currents for motoringe.) ideal currents for generating For 120 0 wide magnets delta connection is used while 180 0 wide magnets require star connection. To reduce the fringing (leakage) flux between neighboring magnets their span is 150 0 - 160 0.

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Electric Drives12 Returning to the linear phase flux linkage variation with rotor position (figure 11.5.b) we have: (11.5) The maximum flux linkage per phase PM is: (11.6) The phase e.m.f., E a, is: (11.7) where r is the electrical angular speed. As two phases conduct at any time the ideal torque T e is constant: (11.8) Between any two current instantaneous commutation the phase current is constant (i d = i dc ) and thus the voltage equation is: (11.9)

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Electric Drives13 (11.10) With (11.11) The ideal speed - torque curve is linear (figure 11.6) like for a d.c. brush PM motor. Figure 11.6. Ideal speed - torque curves of BLDC

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Electric Drives14 11.3.2. The rectangular current control system In general a rectangular current control system contains the BLDC motor, the PWM inverter, the speed and current controllers and the position (speed) sensors (or estimators, for sensorless control) and the current sensors (figure 11.7). Figure 11.7. Rectangular current control of BLDC

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Electric Drives15 The currents sequence, produced through inverter adequate control, with 120 0 current waveforms in figure 11.8 shows also the position of the 6 elements of the proximity sensors with respect to the axis of the phase a for a zero advance angle a = 0. Figure 11.8. a.) Current sequencing b.) phase connection

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Electric Drives16 With two phases conducting the stator active m.m.f. is on from 60 0 to 120 0 with respect to the rotor position. The ideal voltage vector (figure 11.8) also jumps 60 0 for any phase commutation in the inverter. Each phase is on 120 0 out of 180 0 for the 120 0 conducting strategy. To reverse the speed the addresses (IGBTs) of the proximity sensor elements action are shifted by 180 0 (P(a+) P(a-); P(b+) P(b-); P(c+) P(c-)). The proximity sensor has been located for zero advance angle to provide similar performance for direct and reverse motion. However, through electronic means, the advance angle may be increased as speed increases to reduce the peak PM flux in the stator phase and thus produce more torque, for limited voltage, at high speeds. Using the same hardware we may also provide for 180 0 conduction conditions, at high speeds, when all three phases conduct at any time.

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Electric Drives17 11.3.3. The hysteresis current controller Figure 11.9. Current chopping

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Electric Drives18 Figure 11.10. Conduction of phases a and b a.) on - timeb.) off - time

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Electric Drives19 During the on - time, figure 11.10.a, the a+b- equation is: (11.12) (11.13) Note that if the turning on is advanced by a part of the on time E a - E b = 0 and thus a faster current increase is possible. The solution of equation (11.12) is: (11.14) To allow for current rising V d > E 0. Above a certain speed V d < E 0 and thus current chopping is not feasable any more. The current waveform contains in this case a single on - off pulse triggered by the proximity sensor (estimator).

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Electric Drives20 During the off time (diodes D1 and D4 conducting, in figure 11.10.b) the voltage equation is: (11.15) V c0 is the capacitor voltage at the end of on time (11.15), or at the beginning of off - time. The solution of (11.15) with t = t - t on is: (11.16) with(11.17) The torque T e (t) expression is: (11.18) So if the e.m.f. is constant in time the electromagnetic torque reproduces the current pulsations between i min and i max.

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Electric Drives21 Example 11.1. A BLDC motor is fed through a PWM inverter from a 300V d.c. source (V d = 300V). Rectangular current control is performed at an electrical speed r = 2 10rad/s. The no load line voltage at r is E 0 = 48V = const. The cyclic inductance L s = 0.5mH, r s = 0.1 and the stator winding has q = 1 slot per pole per phase and two poles (2p = 2). The filter capacitor C f = 10mF, the current chopping frequency f c = 1.25kHz and t on / t off = 5/3. Determine: a.) the minimum and maximum values of current (i min, i max ) during current chopping; b.) calculate the torque expression and plot it. Solution: a.) To find i min and i max we have to use equation (11.14) for i(t) = i max and t = t on and (11.16) for i(t) = i min and t = t off. (11.19) Notice that t on = 0.5ms and t off = 0.3ms (11.20)

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Electric Drives22 Also from (11.17): (11.21) (11.22) (11.23) (11.24) From (11.19) and (11.24) we may calculate I max and I min : (11.25) The average current. b.) The average torque T av is: (11.26)

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Electric Drives23 The instantaneous torque (11.18) includes current pulsations (as from (11.14) and (11.16), figure 11.11). Figure 11.11. Torque pulsations due to current chopping only It should be noticed that the chopping frequency is low for the chosen speed (E 0 << V d ) and thus the current and torque pulsations are large. Increasing the chopping frequency will reduce these pulsations. To keep the current error band 2(I max - I min ) whithin reasonable limits the chopping frequency should vary with speed (higher at lower speeds and lower at medium speeds) Though the high frequency torque pulsations due to current chopping are not followed by the motor speed, due to the much larger mechanical time constant, they produce flux density pulsations and, thus, notable additional core and copper losses (only the average current I 0 is, in fact, useful).

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Electric Drives24 11.3.4. Practical performance So far the phase commutation transients - current overlapping - have been neglected. They however introduce notable torque pulsations at 6 r frequency (figure 11.12) much lower than those due to current chopping. To account for them complete simulation or testing is required [3]. Figure 11.12. Torque pulsations due to phase commutation

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Electric Drives25 Special measures are required to reduce the cogging torque to less than 2 - 5% of rated torque for high performance drives. While at low speeds current chopping is feasable at high speeds one current pulse remains (figure 11.13). The current controller gets saturated and the required current is not reached. Figure 11.13. Current waveform at high speeds As the advance angle is zero ( a = 0) there is a delay in installing the current and thus, as the e.m.f. is in phase with the reference current, a further reduction in torque occurs.

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Electric Drives26 11.3.5. Extending the torque - speed domain Extending the torque - speed domain may be obtained (for a given drive) by advancing the phase commutation time by an angle a dependent on speed. This phase advancing allows fast current rise before the occurence of the e.m.f. (assuming a PM span angle PM < 150 0 - 160 0 ) An approximate way to estimate the advance angle required a, for 120 0 conduction, may be based on linear current rise [4] to the value I: (11.27) where n - is the rotor speed in rps. Torque at even higher speeds may be obtained by switching from 120 0 to 180 0 current conduction (three phase working at any time). The current waveform changes, especially with advancing angle (figure 11.14). This time the e.m.f. is considered trapezoidal, that is close to reality. The advancing angle a may be, for high speeds, calculated assuming sinusoidal e.m.f. [4] and current variation: (11.28)

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Electric Drives27 Figure 11.14. 180 0 conducting with advancing angle at high speed It has been demonstrated [3,4] that 120 0 conduction is profitable at low to base speeds while 180 0 conduction with advancing angle is profitable for high speeds (figure 11.15). Figure 11.15. Torque - speed curves for various advancing angle a A smooth transition between 120 0 and 180 0 conduction is required to fully exploit the torque - speed capabilities of brushless d.c. motor drives.

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Electric Drives28 Example 11.2. Digital simulation: a brushless d.c. motor drive We will present here the simulation results on a permanent magnet brushless DC motor drive (BLDC). The motor equations (see equation (10.6) in chapter 10) are: (11.29) with(11.30) L a, M self and mutual inductances. Introducing the null point of d.c. link (0), the phase voltages are: (11.31) For star connection of phases: (11.32) Here n represents the star connection point of stator windings. V a0, V b0, V c0 could easily be related to the d.c. link voltage and inverter switching state.

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Electric Drives29 The simulation of this drive was implemented in MATLAB - SIMULINK. The motor model was integrated in a block (PM_SM). The changing of motor parameters for different simulations is as simple as possible. After clicking on this block, a dialog box appears and you can change them by modifying their default values. The drive system consists of a PI speed controller (Ki = 20, Ti = 0.05s), a reference current calculation block, hysteresis controller and motor blocks. The study examines the system behavior for starting, load perturbation and speed reversal. The motor operates at desired speed and the phase currents are regulated within a hysteresis band around the reference currents (as functions of rotor position). The integration step (50 s) can be modified from the Simulinks Simulation / Parameters. To find out the structure of each block presented above unmask it (Options/Unmask). Each masked block contains a short help describing that block (inputs / outputs / parameters).

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Electric Drives30 Figure 11.16. The BLDC rectangular current - drive controller

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Electric Drives31 Figure 11.17. The BLDC motor block diagram

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Electric Drives32 The drive and motor used for this simulation have the following parameters: V dc = 220V, 2p = 2, R s = 1, L s = 0.02H, M = -0.006667H, J = 0.005kgm 2, K = 0.763, hb (hysteresis band) = 0.2. The following figures represent the speed (figure 11.18), torque (figure 11.19) and current (figure 11.20) responses, and e.m.f. waveform (figure 11.21), for a starting process, loading (6Nm) at 0.2s, unloading at 0.4s, reversal at 0.5s and loading (6Nm) again at 0.8s.

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Electric Drives33 Figure 11.18. Speed transient response

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Electric Drives34 Figure 11.19. Torque response

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Electric Drives35 Figure 11.20.Current waveform (i a )

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Electric Drives36 Figure 11.21.Induced voltage waveform (e a )

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Electric Drives37 11.4. VECTOR (SINUSOIDAL) CONTROL The vector (sinusoidal) control is applied both for surface PM and interior PM rotors and distributed stator windings (q 2). According to chapter 10 the torque T e expression is: (11.33) The space vector equation (chapter 10) is: (11.34) (11.35) (11.36)

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Electric Drives38 In general L d L q and thus the second (reactive) torque component is positive only if i d 0. For i d 0 the PM flux is diminished by the i d m.m.f. (figure 11.22) but always the PM prevails to avoid complete PM demagnetization. For vector control, in general, the rotor does not have a damper cage and thus there is no rotor circuit, and, as a direct consequence, no current decoupling network is necessary, in contrast to IM case. Figure 11.22. Space vector diagram of PM - SM with negative i d (i d < 0) In vector control the PM - SM is controlled in d - q coordinates and then the reference d - q currents (or voltages) are transformed into stator coordinates through the vector rotator (inverse Park transformation) to be then realized by PWM in the inverter.

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Electric Drives39 11.4.1. Optimum i d - i q relationship Since the current decoupling network is missing and the drive command variable is the reference torque T e * as required by a speed (speed and position) controller, we should now choose i d * and i q * from the torque equation (11.33). Evidently we need one more equation. This additional information may be obtained through an optimisation criterion such as: maximum torque per current, maximum torque per flux, maximum efficiency etc. As at low speeds the drive is current limited and above base speed b it is flux limited, we may use these two criteria combined for a high performance drive. The maximum torque / ampere criterion is applied by using (11.34) and: (11.37) (11.38) to find: (11.39) For L d = L q, starting with (11.33), it follows that: (11.40)

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Electric Drives40 On the other hand, for maximum torque per flux: (11.41) Proceeding as above we obtain a new relationship between and : (11.42) where: (11.43) For L d = L q (11.41) becomes (starting all over with (11.33)): (11.44) Equation (11.44) signifies complete cancellation of PM flux. The PM is still not completely demagnetized as part of the L d i d is leakage flux which does not flow through the PM. Also notice that constant current i s means a circle in the i d - i q plane and constant stator flux s means an ellipse.

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Electric Drives41 We may now represent (11.38) and (11.40) and (11.41) and (11.42) as in figure 11.23. Figure 11.23. i d - i q a.) for given current i s * b.) for given flux s *

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Electric Drives42 So, for each value of reference torque T e *, according to one of the two optimisation criteria, unique values of i d * and i q * are obtained provided the current limit i s * and the flux limit s * are not surpassed. Notice that the flux limit is related to speed r (through 11.43). So, in fact, the torque T e * is limited with respect to stator current and flux (speed) (figure 11.23.a). (11.45) (11.46) T e * limit is dependent on speed (figure 11.24.b). Figure 11.24. i d - i q optimum relationship and torque and flux limits with speed

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Electric Drives43 Example 11.3. A PM - SM with V sn = 120 V, i sn = 20A, l d = 0.6p.u., no load voltage e 0 = 0.7p.u., l q = 1.2p.u., n n = 15rps, p = 1 (pole pairs), r s = 0, is driven at a current of 10A up to base speed n b according to the criterion of maximum torque per current. Calculate the torque T ei up to base speed and the base speed b. Solution: First we have to calculate l d, l q, e 0 in absolute units L d, L q, E o : (11.47) (11.48) (11.49) The PM flux PM is: (11.50)

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Electric Drives44 Making use of (11.39) with (11.50): (11.51) (11.52) (11.53) and the torque: (11.54)

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Electric Drives45 Now we have to find the maximum speed for which this torque may be produced: (11.55) (11.56) Consequently for rated voltage (maximum inverter voltage) the base speed b is: (11.57) (11.58) In a similar way we may proceed above base speed using the maximum torque / flux criterion.

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Electric Drives46 11.4.2. The indirect vector current control Indirect vector current control means, in fact, making use of precalculated and ((11.45 - (11.46)) to produce the reference d -q currents i d *, i q *. Then, with Park transformation, the reference phase current controllers are used to produce PWM in the inverter. A rotor position sensor is required for position and speed feedback (11.25). Figure 11.25. Indirect vector current control of PM - SM

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Electric Drives47 As for the IMs the a.c. controllers may be replaced by d.c. current controllers (in rotor coordinates) to improve the performance at high speeds, especially. Though, in principle, direct vector control is possible, it is hardly practical unless the drive is sensorless. 11.4.3. Indirect voltage and current vector control As expected, the vector current control does not account for the e.m.f. effect of slowing down the current transients with increasing speed. This problem may be solved by using the stator voltage equation for voltage decoupling: (11.59) (11.60) The d.c. current controllers may replace the first terms in (11.59) - (11.60) and thus only the motion induced voltages are feedforwarded. Then PWM is performed open loop to produce the phase voltages at the motor terminals through the PWM inverter (figure 11.26).

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Electric Drives48 Figure 11.26. Indirect voltage and current vector control of PMSM with dq (dc) current controllers As expected the open loop (voltage) PWM has to observe the voltage limit: (11.61)

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Electric Drives49 At low speeds the current controllers prevail while at high speeds the voltage decoupler takes over. Results obtained with such a method [5], based only on f di, f qi (maximum torque per current criterion) proved a remarkable enlargement of the torque - speed envelope (figure 11.27). Figure 11.27. Torque - speed envelope for i d = 0 and for voltage and current control

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Electric Drives50 11.4.4. Fast response PM - SM drives: surface PM rotor motors with predictive control Interior PM - SM drives have been so far treated for vector control. As L q is rather high in such motors, at least i q response is rather slow in comparison with IM drives. However for surface PM rotor PM - SMs, L d = L q = L s is small and thus fast current (and torque) response may be obtained. The current increment i s is: (11.62) This classical method [7] has produced spectacular results with position sensors (encoders) - figure 11.28. Figure 11.28. Speed responses at low speed The low speed / time linearity [7] is obtained also by using special measures to reduce all torque pulsations to less than 1.5% of rated torque.

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Electric Drives51 Example 11.4. Digital simulation: Indirect vector a.c. current control For the PM - SM motor in example 11.3, making use of the indirect vector current control system of figure 11.25, perform digital simulations. The simulation of this drive was implemented in MATLAB - SIMULINK. The motor model was integrated in a block (PM_SM) (figures 11.29, 11.30, 11.31). Figure 11.29. i d *, i q * referencers

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Electric Drives52 Figure 11.30. A.c. current controllers

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Electric Drives53 Figure 11.31. PM - SM block diagram

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Electric Drives54 The motor used for this simulation has the following parameters: P n = 900W, U n = 220V, 2p = 4, n = 1700rpm; I n = 3A, PM = 0.272Wb, R s = 4.3, L d = 0.027H, L q = 0.067H, J = 0.00179kgm 2.The following figures represent the speed (figure 11.32), torque (figure 11.33) and current waveform (figure 11.34), for a starting process and load torque (2Nm) applied at 0.4s. Figure 11.32. Speed transient response

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Electric Drives55 Figure 11.33. Torque response

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Electric Drives56 Figure 11.34.Current waveform (i a ) under steady state

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Electric Drives57 The d - q reference current relationships to torque for maximum torque per current are given in figure 11.35. Figure 11.35.

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Electric Drives58 Example 11.5. Digital simulation: indirect vector d.c. current control of PM - SM The simulation of this drive (figure 11.36) was implemented in MATLAB - SIMULINK simulation program. The motor model was integrated in the PM_SM block (figure 11.37). The motor used for this simulation has the following parameters: P n = 900W, U n = 220V, 2p = 4, n = 1700rpm; I n = 3A, PM = 0.272Wb, R s = 4.3, L d = 0.027H, L q = 0.067H, J = 0.000179kgm 2.

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Electric Drives59 Figure 11.36. The indirect vector d.c. current controller

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Electric Drives60 Figure 11.37. PM - SM model

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Electric Drives61 Figure 11.38. i d - i q reference currents versus torque for maximum torque / current criterion

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Electric Drives62 The following figures represent the speed (figure 11.39), torque (figure 11.40) and d - q current responses (figure 11.41, 11.42), for a starting process and load torque (2Nm) applied at 0.4s. Figure 11.39. Speed transient response

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Electric Drives63 Figure 11.40. Torque response

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Electric Drives64 Figure 11.41.Current waveform (i d ) under steady state

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Electric Drives65 Figure 11.42.Current waveform (i q ) under steady state

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