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SOC Design: From System to Transistor

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1 SOC Design: From System to Transistor
Zoran Stamenković

2 Outline Modeling Systems Simulation and Verification
Analog Integrated Circuits Digital Integrated Circuits Embedded Memories Logic Synthesis Design for Testability Layout Generation Design for Manufacturability SOC Example

3 Modeling Systems Domains and Levels ESL Design Basics of HDL
Gate Modeling Delay Modeling Power Modeling Effects of Parasitics Logic Optimization

4 Domains and Levels Open Systems Interconnection (OSI) model of network communication Local area network (LAN) technologies are defined by standards that describe unique functions at both the Physical and the Data Link layers

5 Domains and Levels 802.11 Wireless LAN modem
Modulates outgoing digital signals from a computer or other digital device to an analogue (radio) signal Demodulates the incoming analogue (radio) signal and converts it to a digital signal for the digital device

6 MIMO and MIMAX WLAN Modems
Domains and Levels MIMO and MIMAX WLAN Modems Signal processing performed in the digital baseband Signal processing performed in the analogue RF domain Number of the digital basebands reduced to a single one

7 Domains and Levels Structural Behavioral Analysis Synthesis Refinement
Physical Analysis Synthesis Generation Extraction Refinement Abstraction

8 Behavioral Domain Structural Behavioral Algorithm (behavioral)
Physical Algorithm (behavioral) Register-Transfer Language Boolean Equations Differential Equations

9 Processor-Memory System
Structural Domain Behavioral Structural Physical Processor-Memory System Registers Gates Transistors

10 Physical Domain Behavioral Structural Physical Polygons Sticks
Standard Cells Floor Plan

11 Electronic System Level Design
The point of a system level model is to capture the intent of the design Design does exactly what it is defined to do, and the model is the definition of what the design does It allows software developers to test their code on a working model The value of system level modeling is in helping us to understand the implications of our intent To explore responses to the stimulus in an useful way ESL Languages UML, SystemC, SystemVerilog ESL Verification “No amount of experimentation can ever prove me right; a single experiment can prove me wrong” – Albert Einstein The system level testbench languages and methodologies that exist today are woefully inadequate If one tries to capture enough information in ESL to verify RTL, then one might as well write RTL

12 Electronic System Level Design
The environment that provides models of memories, connectors, and queues that can be interconnected with configured processors into an overall system model Processor and device interfaces are at the transaction level Transaction-level modeling requests for SOC architecture assembly and simulation tools If RTL IP blocks present, HW/SW co-verification tools needed Xtensa Processor Generator Use standard ASIC/COT design techniques and libraries for any IC fabrication process Complete Hardware Design Source pre-verified RTL, EDA scripts, test suite Customized Software Tools C/C++ compiler Debuggers Simulators RTOSes Processor Extensions int main( ) { int i; short c[100]; for (i=0;i<N/2;i++) ANSI C/C++ Code XPRES Compiler Configuration Select from menu Add instruction description (TIE) Automatic instruction discovery (XPRES)

13 Electronic System Level Design
Auto-generated XTMP model based on memory maps Specify chip-level memory maps for shared/private memories Place interrupt and reset vectors Assign code/data to distributed memories

14 Hardware Description Languages
Motivation for HDL Increased hardware complexity Design space exploration Inexpensive alternative to prototyping General features Support for describing circuit connectivity High-level programming language support for describing behavior Support for timing information (constraints, etc.) Support for concurrency VHDL IEEE Standard IEEE Standard Extension VHDL-AMS-1999 Verilog IEEE Standard IEEE Standard

15 Modeling Interfaces Entity (VHDL) or Module (Verilog) declaration Describes the input/output ports of a module entity reg3 is port ( d0, d1, d2, en, clk : in bit; q0, q1, q2 : out bit ); end; name port names port mode (direction) port type (VHDL only) reserved words punctuation module reg3 ( d0, d1, d2, en, clk, q0, q1, q2 ); input d0, d1, d2, en, clk; output q0, q1, q2; endmodule VHDL Verilog

16 Modeling Behavior Architecture Body (VHDL)
Describes an implementation of an entity May be several per entity Module (Verilog) Is unique Behavioral Architecture Describes the algorithm performed by the module Contains Procedural Statements, each containing Sequential Statements, including Assignment Statements and Wait Statements

17 Behavior Example VHDL Verilog
entity reg3 is port ( d0, d1, d2, en, clk : in bit; q0, q1, q2 : out bit ); end; architecture behav of reg3 is begin process ( d0, d1, d2, en, clk ) begin if en = '1' and clk = '1' then q0 <= d0 after 5 ns; q1 <= d1 after 5 ns; q2 <= d2 after 5 ns; end if; end process; end; `timescale 1ns/10ps module reg3 ( d0, d1, d2, en, clk, q0, q1, q2 ); input d0, d1, d2, en, clk; output q0, q1, q2; reg q0, q1, q2; ( d0 or d1 or d2 or en or clk ) if ( en & clk ) begin q0 <= #5 d0; q1 <= #5 d1; q2 <= #5 d2; end endmodule VHDL Verilog

18 Modeling Structure Structural Architecture
Implements the module as a composition of components Contains Signal Declarations (entity ports are also signals) Declare internal connections Component Instances Instantiate previously declared entity/architecture pairs Port Maps in component instances Connect signals to component ports Wait Statements Suspend a process or procedure

19 Structure Example d0 d1 d2 q0 q1 q2 bit0 d _latch clk q bit1 bit2 int
en gate and2 a b y

20 Structure Example Verilog VHDL
entity d_latch is port ( d, clk : in bit; q : out bit ); end; architecture basic of d_latch is begin process ( d, clk ) begin if clk = ‘1’ then q <= d after 5 ns; end if; end process; end; entity and2 is port ( a, b : in bit; y : out bit ); end; architecture basic of and2 is begin process ( a, b ) begin y <= a and b after 5 ns; end process; `timescale 1ns/10ps module d_latch ( d, clk, q ); input d, clk; output q; reg q; ( d or clk ) if ( clk ) begin q <= #5 d; end endmodule module and2 ( a, b, y ); input a, b; output y; reg y; ( a or b ) y <= #5 ( a & b ); VHDL Verilog

21 Structure Example Verilog VHDL
entity reg3 is port ( d0, d1, d2, en, clk : in bit; q0, q1, q2 : out bit ); end; architecture struct of reg3 is component d_latch port ( d, clk : in bit; q : out bit ); end component; component and2 port ( a, b : in bit; y : out bit ); end component; signal int_clk : bit; begin bit0 : d_latch port map ( d0, int_clk, q0 ); bit1 : d_latch port map ( d1, int_clk, q1 ); bit2 : d_latch port map ( d2, int_clk, q2 ); gate : and2 port map ( en, clk, int_clk ); end; module reg3 ( d0, d1, d2, en, clk, q0, q1, q2 ); input d0, d1, d2, en, clk; output q0, q1, q2; wire int_clk; d_latch bit0 ( d0, int_clk, q0 ); d_latch bit1 ( d1, int_clk, q1 ); d_latch bit2 ( d2, int_clk, q2 ); and2 gate ( en, clk, int_clk ); endmodule VHDL Verilog

22 Mixing Behavior and Structure
An architecture can contain both behavioral and structural parts Process Statements and Component Instances Collectively called Concurrent Statements Processes can read and assign to signals Example: Register-transfer-language model Data-path described structurally Control section described behaviorally

23 Mixed Example

24 Mixed Example entity multiplier is port ( clk, reset : in bit; multiplicand, multiplier : in integer; product : out integer ); end; architecture mixed of multiplier is signal partial_product, full_product : integer; signal arith_control, result_en, mult_bit, mult_load : bit; begin arith_unit : entity work.shift_adder(behavior) port map ( addend => multiplicand, augend => full_product, sum => partial_product, add_control => arith_control ); result : entity work.reg(behavior) port map ( d => partial_product, q => full_product, en => result_en, reset => reset ); ...

25 Mixed Example multiplier_sr : entity work.shift_reg(behavior) port map ( d => multiplier, q => mult_bit, load => mult_load, clk => clk ); product <= full_product; control_section : process is -- variable declarations for control_section -- … begin -- sequential statements to assign values to control signals -- … wait on clk, reset; end process control_section; end;

26 Logic Functions Function
f = a’b + ab’: a is a variable, a and a’ are literals, ab’ is a term Irredundant Function No literal can be removed without changing its value Implementing logic functions is non-trivial No logic gates in the library for all logic expressions A logic expression may map into gates that consume a lot of area, time, or power A set of functions f1, f2, ... is complete if every Boolean function can be generated by a combination of the functions from the set NAND is a complete set NOR is a complete set AND and OR are not complete Transmission gates are not complete Incomplete set of logic gates No way to design arbitrary logic 1

27 Inverter a out + c a 8

28 Inverter 8

29 Switches Complementary switch produces full-supply voltages for both logic 0 and logic 1 n-type transistor conducts logic 0 p-type transistor conducts logic 1

30 NAND Gate b + a out VDD GND tub ties 9

31 NOR Gate b a out VDD GND tub ties 10

32 AOI/OAI Gates invert out = [ab+c]’ or and AOI = and/or/invert
OAI = or/and/invert Implement larger functions Pull-up and pull-down networks are compact Smaller area, higher speed than NAND/NOR network equivalents AOI312 And 3 inputs And 1 input (dummy) And 2 inputs Or together these terms Invert and or invert out = [ab+c]’ 11

33 logic 1 unknown logic 0 VDD VH VL VSS Logic Levels
Solid logic 0/1 defined by VSS/VDD Inner bounds of logic values VL/VH are not directly determined by circuit properties, as in some other logic families logic 1 logic 0 unknown VDD VSS VH VL Levels at output of one gate must be sufficient to drive next gate

34 Inverter Transfer Curve
Choose threshold voltages at points where slope of transfer curve is -1 Inverter has High gain between VIL and VIH points Low gain at outer regions of transfer curve Note that logic 0 and 1 regions are not equally sized In this case, high pull-up resistance leads to smaller logic 1 range Noise margins are VDD-VIH and VIL-VSS Noise must exceed noise margin to make second gate produce wrong output

35 Inverter Delay Only one transistor is on at the time
Rise time (pull-up on) Fall time (pull-up off) Resistor model of transistor Ignores saturation region Mischaracterizes linear region Gives acceptable results

36 RC Model for Delay Delay
Time required for gate’s output to reach 50% of final value Transition time Time required for gate’s output to reach 10% (logic 0) or 90% (logic 1) of final value Gate delay based on RC time constant Vout(t) = VDD exp{-t/[(Rn+RL)CL]} td = 0.69 RnCL tf = 2.3 RnCL 0.5 mm process Rn = 3.9 kW CL = 0.68 fF td = 0.69 x 3.9 x .68E-15 = 1.8 ps tf = 2.3 x 3.9 x .68E-15 = 6.1 ps For pull-up time, use pull-up resistance Current source model (in power/delay studies) tf = CL (VDD-VSS)/[0.5 k’ (W/L) (VDD-VSS -Vt)2] Fitted model Fit curve to measured circuit characteristics

37 Step Input (VGS = VDD) Approximation

38 Body Effect Source voltage of gates in middle of network may not equal substrate voltage Difference between source and substrate voltages causes body effect Source above VSS Early arriving signal To minimize body effect Put early arriving signals at transistors closest to power supply

39 Power Consumption Clock frequency f = 1/t Energy E = CL(VDD - VSS)2
E x f = f CL(VDD - VSS)2 Almost all power consumption comes from switching behavior A single cycle requires one charge and one discharge of capacitor Static power dissipation Comes from leakage currents Surprising result Resistance of the pull-up/pull-down transistor drops out of energy calculation Power consumption is independent of the sizes of the pull-up and pull-down transistors Static CMOS power-delay product is independent of frequency Voltage scaling depends on this fact

40 Effects of Parasitics Capacitance on power supply is not bad
Can be good in absence of inductance Resistance slows down static gates May cause pseudo-nMOS circuits to fail Increasing capacitance/resistance Reduces input slope Resistance near source is more damaging It must charge more capacitance

41 Optimal Sizing Sometimes, large loads must be driven
Off-chip or by long wires on-chip Sizing up the driver transistors only pushes back the problem Driver now presents larger capacitance to earlier stage Use a chain of inverters Each stage has transistors larger than previous stage a is the driver size ratio, Cbig/Cd= an, ln(Cbig/Cd) = n lna Minimize total delay through the driver chain ttot = ln(Cbig/Cd)(a/lna)td Optimal driver size ratio is aopt = e Optimal number of stages is nopt = ln(Cbig/Cd)

42 Driving Large Fan-Out Fan-out adds capacitance
Increase sizes of driver transistors Must take into account rules for driving large loads Add intermediate buffers This may require/allow restructuring of the logic

43 Path Delay Network delay is measured over paths through network
Can trace a causality chain from inputs to worst-case output Critical path creates longest delay Can trace transitions which cause delays that are elements of the critical path delay To reduce circuit delay, speed up the critical path Reducing delay off the path doesn’t help There may be more than one path of the same delay Must speed up all equivalent paths to speed up circuit

44 False Paths Logic gates are not simple nodes
Some input changes don’t cause output changes A false path is a path which cannot be exercised due to Boolean gate conditions False paths cause pessimistic delay estimates

45 Logic Transformations
Rewrite by using sub-expressions Logic rewrites may affect gate placement Flattening logic Increases gate fan-in Logic synthesis programs Transform Boolean expressions into logic gate networks in a particular library Shallow Logic Deep Logic

46 Logic Optimization Optimization goals
Minimize area, meet delay constraint Technology-independent optimization Works on Boolean expression equivalent Estimates size based on number of literals Uses factorization, resubstitution, minimization, etc. Uses simple delay models Technology-dependent optimization Maps Boolean expressions into a particular cell library May perform some optimizations on addition to simple mapping Allows more accurate delay models

47 Simulation and Verification

48 Simulation Simulation
Tests the functionality of a design’s elaborated model Needs a test bench and a simulation tool Advances in discrete time steps Test Bench Includes an instance of the design under test Applies sequences of test values to inputs Monitors signal values on outputs using simulator Simulation Tools NCSIM (Cadence) VSIM (Mentor Graphics) VCS (Synopsys)

49 Event-Driven Simulation
Event-driven simulation is designed for digital circuit characteristics Small number of signal values Relatively sparse activity over time Event-driven simulators try to update only those signals which change in order to reduce CPU time requirements An event is a change in a signal value A time-wheel is a queue of events Simulator traces structure of circuit to determine causality of events Event at input of one gate may cause new event at gate’s output

50 Switch Simulation A C D B
Special type of event-driven simulation optimized for MOS transistors Treats the transistor as a switch Takes capacitance into account to model charge sharing Can also be enhanced to model the transistor as a resistive switch A B C D

51 Test Bench Example entity test_bench is end;
architecture test_reg3 of test_bench is signal d0, d1, d2, en, clk, q0, q1, q2 : bit; begin dut : entity work.reg3(behav) port map ( d0, d1, d2, en, clk, q0, q1, q2 ); stimulus : process is begin d0 <= ’1’; d1 <= ’1’; d2 <= ’1’; wait for 20 ns; en <= ’0’; clk <= ’0’; wait for 20 ns; en <= ’1’; wait for 20 ns; clk <= ’1’; wait for 20 ns; d0 <= ’0’; d1 <= ’0’; d2 <= ’0’; wait for 20 ns; … wait; end process stimulus; end;

52 Verification To test a refinement of a design
Low-level structural model must be functionally the same as a corresponding behavioral model To include two instances of a design in the test bench To stimulate both with same test values on inputs To compare values of outputs for equality To take account of timing differences Zero delay Unit delay Gate delay RC delay

53 Verification Example architecture regression of test_bench is
signal d0, d1, d2, d3, en, clk : bit; signal q0a, q1a, q2a, q3a, q0b, q1b, q2b, q3b : bit; begin dut_a : entity work.reg4(struct) port map ( d0, d1, d2, d3, en, clk, q0a, q1a, q2a, q3a ); dut_b : entity work.reg4(behav) port map ( d0, d1, d2, d3, en, clk, q0b, q1b, q2b, q3b ); stimulus : process is begin d0 <= ’1’; d1 <= ’1’; d2 <= ’1’; d3 <= ’1’; wait for 20 ns; en <= ’0’; clk <= ’0’; wait for 20 ns; en <= ’1’; wait for 20 ns; clk <= ’1’; wait for 20 ns; … wait; end process stimulus; ...

54 … Verification Example
verify : process is begin wait for 10 ns; assert q0a = q0b and q1a = q1b and q2a = q2b and q3a = q3b report ”implementations have different outputs” severity error; wait on d0, d1, d2, d3, en, clk; end process verify; end architecture regression;

55 Annotation Standard Delay Format (SDF) annotation
Design timing is stored in an SDF file Used to iteratively improve design Updates a more-abstract design with information from later design stages Annotation of logic schematic with extracted parasitic resistances and capacitances Back annotation requires tools to know more about each other Simulation tools Synthesis tools Layout tools

56 Standard Delay Format (CELL (CELLTYPE "exnor2_1")
(DELAYFILE (SDFVERSION "OVI 1.0") (DESIGN "tcp_1_chip") (DATE "Fri Apr 30 09:48: ") (VENDOR "cdr3synPwcslV225T125") (PROGRAM "Synopsys Design Compiler cmos") (VERSION " ") (DIVIDER /) (VOLTAGE 2.25:2.25:2.25) (PROCESS) (TEMPERATURE :125.00:125.00) (TIMESCALE 1ns) (CELL (CELLTYPE "tcp_1_chip") (INSTANCE) (DELAY (ABSOLUTE (INTERCONNECT U5/x U81/a (0.000:0.000:0.000)) (INTERCONNECT U73/x U74/a (0.000:0.000:0.000)) ... ) (CELL (CELLTYPE "exnor2_1") (INSTANCE i_aes_wr/U_ALG/U6533) (DELAY (ABSOLUTE (IOPATH a x (0.662:1.045:1.045) (0.682:1.076:1.076)) (IOPATH b x (1.379:1.416:1.416) (1.454:1.492:1.492)) ) ... (CELLTYPE "mux2_2") (INSTANCE i_mips/u0/ejt_tap\/pa_addr_reg_next\/bit_00i/U1) (IOPATH d0 x (0.395:0.395:0.395) (0.464:0.464:0.464)) (IOPATH d1 x (0.387:0.403:0.403) (0.447:0.477:0.477)) (IOPATH sl x (1.768:1.781:1.781) (1.879:1.892:1.892))

57 Analog Integrated Circuits
Filters Amplifiers Phase Lock Loop Voltage Control Oscillator Modulator/Demodulator

58 Fairchild Semiconductor μA741 Op-Amp
In 1963, a 26-year-old engineer named Robert Widlar designed the first monolithic op-amp IC, the μA702 Price at the beginning was $300 Fairchild and competitors have sold it in the hundreds of millions Now, for $300 you can get about a thousand of today’s 741 chips

59 Signetics NE555 Timer A simple IC from 1971 that could function as a timer or an oscillator It would become a best seller in analog semiconductors Kitchen appliances Toys Spacecraft A few thousand other things Many billions have been sold

60 Intersil ICL8038 Waveform Generator
A generator of sine, square, triangular, sawtooth, and pulse waveforms from 1983 Countless applications Music synthesizers “Blue boxes” Hundreds of millions sold Intersil discontinued the production in 2002

61 LNA in BiCMOS Technology

62 PLL for a WLAN

63 Oscillator

64 Modulator

65 Digital Integrated Circuits
Adders Multipliers Shifters Carry Units Arithmetic-Logic Units

66 Full Adder Computes one-bit sum and carry si = ai  bi  cin
cout = aibi + aici + bicin Ripple-carry adder: n-bit adder built from full adders Delay of ripple-carry adder goes through all carry bits

67 Combinational Multiplier
multiplicand x multiplier partial product

68 + Array Multiplier xny0 P(2n-1) P(2n-2) x0y0 x1y0 x2y0 x0y1 x1y1 x0y2
Array multiplier is an efficient layout of a combinational multiplier Array multipliers may be pipelined to decrease clock period at the expense of latency xny0 + P(2n-1) P(2n-2) x0y0 x1y0 x2y0 x0y1 x1y1 x0y2 x1y2 P0

69 Wallace Tree Reduces depth of adder chain Built from carry-save adders
Three inputs a, b, c Produces two outputs y, z y + z = a + b + c Carry-save equations yi = parity (ai,bi,ci) zi = majority (ai,bi,ci) At each stage, i numbers are combined to form 2i/3-sums Final adder completes the summation Wiring is more complex

70 Serial-Parallel Multiplier
Used in serial-arithmetic operations Multiplicand can be held in place by register Multiplier is shifted into array

71 data 1 data 2 output n bits Barrel Shifter
Can perform n-bit shifts in a single cycle Accepts 2n data inputs and n control signals, producing n data outputs Selects arbitrary contiguous n bits out of 2n input buts Examples Right shift: data into top, 0 into bottom Left shift: 0 into top, data into bottom Rotate: data into top and bottom data 1 data 2 n bits output

72 Barrel Shifter Two-dimensional array of 2n vertical X n horizontal cells Input data travels diagonally upward Output wires travel horizontally Control signals run vertically Exactly one control signal is set to 1, turning on all transmission gates in that column Large number of cells, but each one is small Delay is large, considering long wires and transmission gates

73 Carry-Lookahead Unit First computes carry propagate and generate
Pi = ai + bi Gi = aibi Computes sum and carry from P and G si = ci  Pi  Gi ci+1 = Gi + Pici Can recursively expand carry formula ci+1 = Gi + Pi(Gi-1 + Pi-1ci-1) ci+1 = Gi + PiGi-1 + PiPi-1 (Gi-2 + Pi-1ci-2) Expanded formula does not depend on intermediate carries Allows carry for each bit to be computed independently

74 Depth-4 Carry-Lookahead Unit
Deepest carry expansion requires gates with large fan-in Large and slow Carry-lookahead unit requires complex wiring between adders and lookahead unit Values must be routed back from lookahead unit to adder

75 Carry-Skip Adder Looks for cases in which carry out of a set of bits is identical to carry in Typically organized into m-bit stages If ai = bi for every bit in stage, then bypass gate sends stage’s carry input directly to carry output

76 Carry-Select Adder Computes two results in parallel, each for different carry input assumptions Uses actual carry in to select correct result Reduces delay to multiplexer

77 Manchester Carry Chain
Precharged carry chain which uses P and G signals Propagate signal connects adjacent carry bits Generate signal discharges carry bit Worst-case discharge path goes through entire carry chain

78 Serial Adder May be used in signal-processing arithmetic where fast computation is important but latency is unimportant LSB control signal clears the carry shift register

79 Arithmetic-Logic Unit
Computes a variety of logical and arithmetic functions based on opcode May offer complete set of functions of two variables or a subset Built around adder, since carry chain determines delay Function block may be used to compute required intermediate signals for a full-function ALU Transmission gates may introduce significant delay

80 Arithmetic-Logic Unit
P and G compute intermediate values from inputs May not correspond to carry lookahead P and G for non-addition functions Add unit is adder of choice Output unit computes from sum, propagate signal

81 Acorn Computers ARM1 Processor
32-bit RISC microprocessor from 1985 The simplicity made all the difference Small, low power, and easy to program ARM architecture has become the dominant embedded processor More than 10 billion ARM cores have been used in all sorts of gadgetry, including the iPhone

82 Computer Cowboys Sh-Boom Processor
Russell Fish and Chuck Moore 1988 found a way to have the processor run its own super fast internal clock while still staying synchronized with the rest of the computer In the years since Sh-Boom’s invention, the speed of processors had by far surpassed that of motherboards, and so practically every maker of computers and consumer electronics wound up using the same solution Since 2006, Patriot Scientific (and Moore) have reaped over US $125 million in licensing fees from Intel, AMD, Sony, Olympus, and others

83 8-bit Microprocessors Microchip Technology PIC16C84 8-bit microcontroller in 1993 Incorporates EEPROM Does not need UV light to be erased as EPROM needs Radiation-hardened RCA CDP bit microprocessor in 1976 One of the first, if not the first, CMOS processors Low power consumption, wide range of operating voltages and military operating temperature range

84 Embedded Memories Read-Only Memory Static Random-Access Memory
Dynamic Random-Access Memory Memory Generators

85 Memory Architecture Address is divided into row and column
Row may contain full word or more than one word Selected row drives/senses bit lines in columns Amplifiers/drivers read/write bit lines

86 Read-Only Memory (ROM)
ROM core is organized as an array of NOR gates Pull-down transistors of NOR determine programming Erasable ROMs require special processing that is not typically available ROMs on digital ICs are generally mask-programmed Placement of pull-downs determines ROM contents

87 Static Random-Access Memory (SRAM)
Core cell uses six-transistor circuit to store value Value is stored symmetrically Both true and complement are stored on cross-coupled transistors SRAM retains value as long as power is applied to circuit Read Precharge bit and bit’ high Set select line high from row decoder One bit line will be pulled down Write Set bit/bit’ to desired (complementary) values Set select line high Drive on bit lines will flip state if necessary

88 SRAM Sense Amplifier Differential pair
Takes advantage of complementarity of bit lines One bit line goes low One arm of diff pair reduces its current, causing compensating increase in current of another arm Sense amp can be cross-coupled to increase speed

89 Dynamic Random-Access Memory (DRAM)
Cell can easily be made with a CMOS digital technology process Dynamic RAM loses value due to charge leakage Must be refreshed Value is stored on gate capacitance of transistor t1 Read read = 1, write = 0, read_data’ is precharged t1 will pull down read_data’ if 1 is stored Write read = 0, write = 1, write_data = value Guard transistor writes value onto gate capacitance Modern commercial DRAMs use one-transistor cell

90 Toshiba NAND Flash Memory
In 1980, Fujio Masuoka recruited four engineers to a project aimed at designing a memory chip that could store lots of data and would be affordable Team came up with a variation of EEPROM that featured a memory cell consisting of a single transistor (at the time, conventional EEPROM needed two transistors per cell) Why is it named “flash”? Because of the chip’s ultrafast erasing capability In 1984 Masuoka presented a paper at the IEEE International Electron Devices Meeting In 1988 Intel developed a type of flash based on NOR logic gates (a 256‑kilobit chip) Toshiba’s first NAND flash (greater storage densities but trickier to manufacture) hit the market in 1989

91 Memory Generators A software tool which can create memories (ROM or RAM blocks) in a range of sizes as needed The customer usually wants a particular number of words (depth) and bits (width) for each memory ordered Each of the final building blocks (physical layout) will be implemented as a stand-alone, densely packed, pitch-matched array Complex layout generators and state-of-the-art logic and circuit design techniques offer Embedded memories of extreme density and performance Each memory generator is a set of various, parameterized generators Layout generator generates an array of custom, pitch-matched leaf cells Schematic generator and Net-lister extracts a net-list used for both layout vs. schematic and functional verification Function and Timing model generators create models for gate level simulation, dynamic/static timing analysis and synthesis Symbol generator generates schematic Critical Path generator is used for both circuit design and timing characterization

92 Logic Synthesis Logic Synthesis Flow Optimization Technology Mapping
Low-Power Techniques

93 Architectural Description Computer-Aided Synthesis
Logic Synthesis Flow Goal is to create a logic gate network which performs a given set of functions Input is Boolean formulae Output is gates implementing Boolean functions Several iterations needed for generation of the optimized gate-level description Logic synthesis Maps onto available gates Restructures for delay, area, testability, power, etc. Automated logic synthesis has enabled Enormous reduction of the time needed for conversion of a design from high-level to gate-level description Saving of designer resources for architectural and RTL descriptions, and optimization of the standard cell library Optimized Net-list SDF File yes Architectural Description RTL Description Computer-Aided Synthesis Design Constraints Constraints Met no Gate-Level Net-list Standard Cell Library

94 High-Level Synthesis Scheduling determines
Number of clock cycles required As-soon-as-possible (ASAP) schedule puts every operation as early in time as possible As-late-as-possible (ALAP) schedule puts every operation as late in schedule as possible Binding determines Area and cycle time Area tradeoffs must consider Shared function units vs. multiplexers and control Delay tradeoffs must consider Cycle time vs. number of cycles

95 Logic Synthesis Phases
Technology-independent optimizations A Boolean network is the main representation of the logic functions Each node can be represented as sum-of-products (or product-of-sums) Functions in the network need not correspond to logic gates Technology mapping (library binding) Design transformation from technology-independent to technology-dependent Technology-dependent optimizations Work in the available set of logic gates

96 Technology-Independent Optimization
Area is estimated by number of literals Literal is true or complement form of a variable Simplification Rewrites a node to reduce the number of literals in it Network restructuring Introduces new nodes for common factors Collapses several nodes into one new node Delay restructuring Changes factorization to reduce path length out1 = k2 + x2’ out2 = k3 + x1 k2 = x1’ x2 x4 + k1 k3 = k1 x4’ k1 = x2 + x3 x1 x2 x3 x4

97 Covers and Cubes x2 1 x1 x3 Function is defined by
1 Function is defined by On-set: set of inputs for which output is 1 Off-set: set of inputs for which output is 0 Don’t-care-set: set of inputs for which output is don’t-care Each way to write a function as a sum-of-products is a cover It covers the on-set A cover is composed of cubes Cubes are product terms that define a subspace cube in the function space

98 Covers and Optimizations
Larger cover x1’ x2’ x3’ + x1 x2’ x3’ + x1’ x2 x3’ + x1 x2 x3 Requires four cubes (12 literals) Smaller cover x2’ x3’ + x1’ x3’ + x1 x2 x3 Requires three cubes (7 literals) x1’ x2 x3’ is covered by two cubes Don’t-cares Can be implemented in either on-set or off-set Provide the greatest opportunities for minimization in many cases Espresso A two-level logic optimizer Expands, makes irredundant and reduces Optimization loop refines cover to reduce its size

99 Factorization Based on division Formulate candidate divisor
Test how it divides into the function If g = f/c, we can use c as an intermediate function Algebraic division Doesn’t take into account Boolean simplification Less expensive then Boolean division Three steps Generate potential common factors and compute literal savings if used Choose factors to substitute into network Restructure the network to use the new factors Algebraic/Boolean division is used to implement first step

100 Technology Mapping Rewrites Boolean network
In terms of available logic functions Optimizes for Area Delay Can be viewed as a pattern matching problem Find pattern match which minimizes area/delay cost Procedure Write Boolean network in canonical NAND form Write each library gate in canonical NAND form Assign cost to each library gate Use dynamic programming to select minimum-cost cover of network by library gates

101 not optimal, but reasonable cuts usually work well
Breaking into Trees not optimal, but reasonable cuts usually work well

102 Mapping Example after three levels of matching

103 after four levels of matching
Mapping Example after four levels of matching

104 Low Power Techniques Architecture-driven supply voltage scaling
Add extra logic to increase parallelism so that system can run at lower frequency Power improvement for n parallel units over Vref Pn(n) = [1 + Ci(n)/nCref + Cx(n)/Cref](V/Vref) Dynamic voltage and frequency scaling Decreased to parts of the circuit where it does not adversely affect the performance Dynamic scaling is regulated by software based on system load Reducing capacitances Parasitic capacitances of the transistors Parasitic capacitances of the wires

105 Low Power Techniques Reducing switching activity
Deactivate the clock to unused registers (clock gating) Deactivate signals if not used (signal gating) Deactivate VDD for unused hardware blocks (power gating) Distributed clocks: Globally Asynchronous Locally Synchronous Eliminating centrally synchronous clocks and utilizing local clocks Distinct local clocks, possibly running at different frequencies

106 Design for Testability
DFT Methods Scan Design Test Pattern Generation Built-In Self-Test

107 Design for Testability Methods
Make the system as testable as possible Keep minimum cost in hardware and testing time Use knowledge of architecture to help in selection of testability points Modify architecture to improve testability DFT for digital circuits Ad-hoc methods Avoid asynchronous feedback Make flip-flops initializable Avoid redundant gates, large fan-in gates and gated clocks Provide test control for difficult-to-control signals Consider ATE requirements (tri-states, etc.) Structured methods Scan Design Built-in self-test (BIST) Boundary scan

108 Scan Design Circuit is designed using pre-specified design rules
Test structure (hardware) is added to the verified design Add a test control (TC) primary input Replace flip-flops by scan flip-flops (SFF) and connect to form one or more shift registers (scan-chains) in the test mode Make input/output of each scan-chain controllable/observable from primary input/primary output Use combinational ATPG to obtain tests for all testable faults in the combinational logic Add shift register tests and convert ATPG tests into scan sequences for use in manufacturing test Full scan is expensive Must roll out and roll in state many times during a set of tests Partial scan selects some registers (not all) for scanability to reduce the chain length Analysis is required to choose which registers are best for scan

109 Scanable Flip-Flop D TC SD CK Q MUX D flip-flop Master latch
Slave latch Normal mode, D selected Scan mode, SD selected Master open Slave open t Logic overhead

110 Level-Sensitive Scanable Flip-Flop
D SD MCK Q D flip-flop Master latch Slave latch SCK TCK Normal mode Scan Logic overhead

111 Scan Structure SFF Combinational logic PI PO SCANOUT TC or TCK SCANIN
Not shown: CK or MCK/SCK feed all SFFs

112 Combinational Test Vectors
PI PO SCANIN SCANOUT S1 S2 N1 N2 TC Sequence length = (ncomb + 1) nsff + ncomb clock periods ncomb = number of combinational vectors nsff = number of scan flip-flops

113 Testing Scan Chain Scan-chain must be tested prior to application of scan test sequences A shift sequence of length nsff+4 in scan mode (TC=0) Produces 00, 01, 11 and 10 transitions in all flip-flops Observes the result at SCANOUT output Total scan test length (ncomb + 2) nsff + ncomb + 4 clock periods Example 2,000 scan flip-flops, 500 comb. vectors, total scan test length ~ 106 clocks Multiple scan-chains reduce test length

114 Testing and Faults Errors are introduced during manufacturing
Testing weeds out infant mortality Varieties of testing Functional testing Performance testing Fault model Possible locations of faults I/O behavior produced by the fault With a fault model, we can test the network for every possible instantiation of that type of fault It is difficult to enumerate all types of manufacturing faults Testing procedure Set inputs Observe output Compare fault-free and observed output

115 Stuck-At-0/1 Faults Logic gate output is always stuck at 0 or 1 independently on input values Correspondence to manufacturing defects depends on logic family Experiments show that 100% stuck-at-0/1 fault coverage corresponds to high overall fault coverage Testing NAND Three ways to test it for stuck-at-0 Only one way to test it for stuck-at-1 Testing NOR Three ways to test it for stuck-at-1 Only one way to test it for stuck-at-0 a b OK SA0 SA1 a b OK SA0 SA1

116 Multiple Test Example Can test both NANDs for stuck-at-0 simultaneously abc = 000 Cannot test both NANDs for stuck-at-1 simultaneously due to inverter Must use two vectors Must also test inverter

117 Stuck-At-Open/Closed Model
Transistors always on/off t1 is stuck open (switch cannot be closed) No path from VDD to output capacitance Testing requires two cycles Must discharge capacitor Try to operate t1 to charge capacitor

118 Combinational Testing Example
Two parts of testing Controlling the inputs of (possibly interior) gates Observing the outputs of (possibly interior) gates Delay faults Gate delay model assumes that all delays are lumped into one gate Path delay model takes into account the delay of a path through network Performance problems Functional problems in some types of circuits

119 Testing Procedure Goal Test gate D for stuck-at-0 fault First step
Justify 0 values on gate inputs Work backward from gate to primary inputs w1 = 0 (A output = 0) i1 = i2 = 1 Observe the fault at a primary output o1 gives different values if D is true/faulty Work forward and backward F’s other input must be 0 to detect true/fault Justify 0 at E’s output In general, may have to propagate fault through multiple levels of logic to primary outputs

120 Redundancy and Testing
Redundant logic can mask faults Testing NOR for SA0 requires setting both inputs to 0 Network topology ensures that one NOR input (for instance b) will always be 1 Function reduces to 0 f = ((a+b)’ + b)’ = (a + b)b’ = 0 Redundant logic can introduce delay faults and other problems

121 Sequential Testing Much harder than combinational testing
Can’t set memory element values directly Must apply sequences To put machine in proper state for test To observe value of test Testing of NAND for stuck-at-1 Set both NAND inputs to 1 Primary input i1 can be controlled directly Lower input is 1 if ps0/ps1 = 1

122 Time-Frame Expansion A model for sequential test
Unroll machine in time A single-stuck-at fault in sequential machine appears to be the multiple-stuck-at fault

123 Test Pattern Generation
Automatic test pattern generator (ATPG) generates a set of test vectors Boolean network (combinational ATPG) Sequential machine (sequential ATPG) D (from Discrepancy) allows us to quickly write fault D value on a node means that good and faulty circuits have different values at that point If a test for a particular fault exists, D-algorithm will find it by an exhaustive search of all sensitized paths Start at the faulty gate Suppose initially a stuck-at fault on gate output “Primitive D-cube of failure” (PDCF) of gate summarizes minimal assignment of input values to highlight fault Propagation D-cube (PDC) has D or D’ on output and on at least one input Summarizes “non-controlling” values for other inputs to allow propagation of D signal

124 PODEM Algorithm PODEM stands for Path-Oriented DEcision Making
Circuit-based, fault-oriented ATPG algorithm Goal Propagate D value to primary outputs Signal values are explicitly assigned at primary inputs only Other values are computed by implication Backtracking means reassigning primary inputs when a contradiction occurs Uses implicit enumeration Uses five values: 0, 1, D, D’, and X Start all values at X In worst case, must examine all possible inputs Can be implemented to run quickly

125 Fault Propagation Example

126 Built-In Self-Test (BIST)
Includes on-chip machine responsible for Generating tests Evaluating correctness of tests Allows many tests to be applied Can’t afford large memory for test results Rely on compression and statistical analysis Uses a linear-feedback shift register (LFSR) to generate a pseudo-random sequence of bit vectors

127 BIST Architecture One LFSR generates test sequence
Another LFSR captures/compresses results Can store a small number of signatures which contain expected compressed results for valid system Usually used for testing memory blocks

128 Layout Generation Layout Generation Flow Design Rules Layout Tools
Standard Cells Floorplanning Placement Routing Clock Tree Pads

129 Layout Generation Flow
Verify Net-list Floorplanning Plan Power Routing Placement Adjust Size Generate Clock Trees Optimize Placement Route Post-layout Net-list SDF, DEF, GDSII Cell Library (LEF, TLF) Design Constraints Library Exchange Format (LEF) files To create a library database (standard cells, I/O cells, and macro blocks) Timing Library Format (TLF) file Timing constraints General Constraints Format (GCF) file Design constraints Verilog net-list To create a design database

130 Layout Generation Flow
Floorplanning To create a core area with rows (or columns) and I/O rows around the core area Power planning and routing To plan, modify and rout power paths, power rings and power stripes Placement An I/O constraints file may be used to place the I/O pads Block placement Cell placement Size adjustment To estimate the die size To resize the design to make it routable

131 Layout Generation Flow
Generating clock trees The clock buffer space and clock net must be defined Generating clock trees is iterative process At this point, the physical net-list differ from the logical (original) net-list Placement optimization To resize gates and insert buffers to correct timing and electrical violations Routing To perform both global and final route on a placed design Verification To check for shorts and design rule violations

132 Design Rules Masks are tools for manufacturing
Manufacturing processes have inherent limitations in accuracy Design rules specify geometry of masks which will provide reasonable yields Design rules are determined by experience MOSIS SCMOS Designed to scale across a wide range of technologies Designed to support multiple vendors Designed for educational use Fairly conservative Lambda () design rules Size of a minimum feature defines  Specifying  particularizes the scalable rules Parasitics are generally not specified in units

133 Wires metal 3 6 metal 2 3 metal 1 pdiff/ndiff poly 2

134 Transistors 2 3 1 5

135 Vias 4 1 2 ... Types of via Metal1/diff Metal1/poly Metal2/metal1
Highest via Cut: 3 x 3 Overlap by metal2: 1 Minimum spacing: 3 Minimum spacing to via1: 2 4 1 2

136 Spacings Diffusion/diffusion 3 Poly/poly 2 Poly/diffusion 1 Via/via
Metal1/metal1 Metal2/metal2 4 Metal3/metal3

137 Overglass Cut in passivation layer Connection for bonding wire
Minimum bonding pad 100 Pad overlap of glass opening 6 Minimum pad spacing to unrelated metal2/3 30 Minimum pad spacing to unrelated metal1, poly, active 15

138 Layout Tools Layout editors are interactive tools
Design rule checkers identify errors on the layout Circuit extractors extract the net-list from the layout Connectivity verification systems (CVS) compare extracted and original net-lists CADENCE Virtuoso’s Layout-versus-Schematic (LVS) tool Standard cell layouts are created from pre-designed cells using the custom routing Silicon Ensemble (CADENCE) Encounter (CADENCE) Physical Compiler (SYNOPSYS)

139 Standard Cell Layout routing area VDD pullups Feedthrough area n tub
Layout made of small cells Gates, flip-flops, etc. Cells are hand-designed Assembly of cells is automatic Cells arranged in rows Wires routed between and through cells Pitch is the height of a cell All cells have same pitch, may have different widths VDD/VSS connections are designed to run through cells A feedthrough area allows wires to be routed over the cell routing area VSS n tub p tub Intra-cell wiring pullups pulldowns pin Feedthrough area VDD

140 Floorplanning Strategy
Floorplanning must take into account Blocks of varying function, size, and shape Space allocation Signal routing Power supply routing Clock distribution

141 Floorplanning Tips Develop a wiring plan
Think about how layers will be used to distribute important wires Draw separate wiring plans for power and clocking These are important design tasks which should be tackled early Sweep small components into larger blocks A floorplan with a single NAND gate in the middle will be hard to work with Design wiring that looks simple If it looks complicated, it is complicated Design planar wiring Planarity is the essence of simplicity Do it where feasible (and where it doesn’t introduce unacceptable delay)

142 Placement Metrics bad placement good placement
Placement of components interacts with routing of wires Quality metrics for layout Area and delay Area and delay determined in part by Wiring How do we judge a placement without wiring? Estimate wire length without actually performing routing bad placement good placement

143 Placement Techniques To construct an initial solution
To improve an existing solution Pairwise interchange is a simple improvement metric Interchange a pair, keep the swap if it helps wire length Heuristic determines which two components to swap Placement by partitioning Works well for components of fairly uniform size Partition net-list to minimize total wire length using min-cut criterion Kernighan-Lin Algorithm Computes min-cut criterion, count total net-cut change Exchanges sets of nodes to perform hill-climbing finding improvements where no single swap will improve the cut Recursively subdivide to determine placement detail

144 Routing channel switchbox Major phases in routing
Global routing assigns nets to routing areas Detailed routing designs the routing areas Net ordering determines quality of result Net ordering is a heuristic Blocks and wiring Blocks divide wiring area into routing channels Large wiring areas may force rearrangement of block placement Channel routing Channel grows in one dimension to accommodate wires Pins generally on only two sides Switchbox routing Box cannot grow in any dimension Pins are on all four sides channel switchbox

145 Routing Channels Tracks form a grid for routing
Spacing between tracks is center-to-center distance between wires Track spacing depends on wire layer used Density (vertical and horizontal) Gives the number of wire segments crossing a vertical/horizontal grid segment Different layers are used for horizontal and vertical wires Horizontal and vertical wires can be routed relatively independently Placement of cells determines placement of pins Pin placement determines difficulty of routing problem

146 Left-Edge Algorithm B A aligned ? A B C
Assumes one horizontal segment per net Sweep pins from left to right Assign horizontal segment to lowest available track Limitations Some combinations of nets require more than one horizontal segment per net (a dog-leg wire) Aligned pins form vertical constraints Wire to lower pin must be on lower track Wire to upper pin must be above lower pin’s wire B A aligned ? A B C

147 Global and Detailed Routing
Global routing Assign wires to paths through channels Don’t worry about exact routing of wires within channel Can estimate channel height using congestion Detailed routing Dog-leg router breaks net into multiple segments as needed Minimize number of dog-leg segments per net to minimize congestion for future nets Use left-edge criterion on each dog-leg segment to fill up the channel

148 Multipoint Nets Multipoint nets are harder to design than two-point nets Rectilinear Steiner tree problem: Find the minimum-length tree interconnecting all net’s points Find additional points (so-called Steiner points) in the plane, if they contribute to a shorter tree length Steiner tree algorithm Compute the spanning tree Optimize the tree length by flipping L-shaped branches Steiner points always have degree three

149 Layout for Low Power Place and route to minimize
Capacitance of nodes with high glitching activity Feed back wiring capacitance values To better estimate power consumption Size wires to be able to handle current Requires designing topology of VDD/VSS networks Keep power network in metal Requires designing planar wiring

150 Clock Delay Clock delay varies with position
Deliver clock to memory elements with acceptable skew Deliver clock edges with acceptable sharpness Clocking network design The greatest challenge in the design of a large chip

151 Clock Distribution Tree
Clocks are generally distributed via wiring trees Use low-resistance interconnect to minimize delay Use multiple drivers to distribute driver requirements Use optimal sizing principles to design buffers Clock lines can create significant crosstalk

152 Pad Architecture Pads are placed on top-layer metal
Provide a place to bond to the package Pads are typically placed around periphery of chip Some advanced packaging systems bond directly to package without bonding wire Some allow pads across entire chip surface Supply power/ground to Each pad Chip core Positions of pads May be determined by pin requirements Distribute power/ground pins as evenly as possible To minimize power distribution problems

153 Input Pad Main purpose is to provide electrostatic discharge (ESD) protection Gate of transistor is very sensitive Can be permanently damaged by high voltage Static electricity in room is sufficient to cause damage Resistor is used in series with pad to limit current caused by voltage spike May use parasitic bipolar transistors to drain away high voltages One for positive pulses Another for negative pulses Must design layout to avoid latch-up

154 Output Pad Don’t need ESD protection
Transistor gates not connected to pad Must be able to drive capacitive pad load and outside load May need voltage level shifting To be compatible with other logic families

155 Three-State Pad Combination of input and output
Controlled by a mode-input on chip Pad includes logic to disconnect output driver when pad is used as an input Must be protected against ESD

156 Design for Manufacturability
Defects and Faults Critical Area Modeling Critical Area Extraction Yield Modeling Yield Control

157 Defects, Faults, and Tests
Design faults Random faults Shorts between lines Breaks of lines Leakage of insulation layers Permanent faults Dynamic faults Transient faults Parametric test Power consumption Input resistance Input/output current Functional test Test signals applied Output observed

158 Defects Statistics Defect density distribution g(D)
Defect size distribution h(X)

159 Defect Density Distribution
g(D) 2D 1

160 Defect Size Distribution

161 Test Structures Sample size Number of measurements Structure size
High and low defect densities Critical dimensions As defect sizes Self-isolation Sensitive on one defect type Measurability Electrical Simple Reliable

162 Long parallel conductors
Critical Area Models Long parallel conductors Real Patterns

163 Critical Area Models

164 Critical Area Extraction
Point defects Lithographic defects

165 Critical Area Extraction
Vertical shorts Shorts Breaks

166 Yield Models

167 Yield Control

168 SOC Example Design Methodology SOC Design Flow An Example

169 Design Methodology Every company has its own design methodology
Methodology depends on Size of chip Design time constraints Cost/performance Available tools Driven by contradictory impulses Customer concerns about cost and performance Forecasts of feasibility of cost and performance Design features, performance, power, etc. May be negotiated at early stages Negotiation at later stages creates problems

170 Design Styles Design styles Full custom logic design (very tedious)
Semi custom or standard cell design Field Programmable Gate Array (FPGA) design Full custom design Most likely for data-paths Least likely for random logic off critical path Standard cell design Application Specific Integrated Circuits (ASIC) FPGA design Prototyping oriented

171 Design Validation Functionality, speed, power consumption, area, etc.
Estimation techniques of circuit performance vary with module Memories may be generated once size is known Data-paths may be estimated from previous design Controllers are hard to estimate without details Clock distribution Layout design rule check Testing Generation of simulation test vectors Generation of scan test vectors (ATPG) Manufacturing test vectors comprise of both

172 SOC Design Flow MPL Library New RTL Designs HDL Top Module Definition
Simulation OK? Logic Synthesis Layout Synthesis Final Chip Layout Test Benches yes Configurable Modules (synthesizable RTL code) Pre-defined Modules (synthesized net-lists, layouts, standard cells) New logic run sufficient? New layout run no Applications System Specification A

173 Applications RF Baseband DLC Wireless Broadband Networks
A highly integrated broadband wireless modem Wireless Internet A new terminal oriented TCP/IP for wireless systems Wireless Sensor Networks A flexible sensor network node architecture for medical applications Mobile Business Engine A specific application processor for highly efficient encryption operations RF Baseband DLC Application Engine Mobile Bus. Protocol Wireless Internet Power Management Test Project Wireless Broadband Network

174 Reusable Modules SOC designs are a mix of
Intellectual Property (IP) blocks Standard functions (in-house developed) Application specific blocks (in-house developed) RTL descriptions, net-lists and layouts Soft-core MIPS32 4KEp Soft-core LEON-2 Hard-core LEON-3 Soft-core IPMS430 UART, GPIO, PCMCIA AMBA, I2C bus Controllers Hardware accelerators SRAM memory generator Hard-core flash

175 Configurable Processor

176 Extensible Processor Customisation addresses three architectural levels Instruction extension: designer specifies its functionality Inclusion/exclusion of predefined blocks: special registers, BIST, etc. Parameterisation: cache size, number of registers, etc. To customise the extensible processor to a specific application It starts with profiling the application using instruction-set simulator To evaluate customisations using retargetable tool generation Retargetable techniques automatically generate compilers and simulators aware of the new or extended instructions Major players in the field Tensilica, Improv Systems, ARC, Coware, and Target Compiler Techn. NEC’s TCP/IP offload engine integrates 10 extensible processors

177 SOC Design Trends “The crisis of complexity” in SOC design
SOC designs do not exploit the possible 100 M transistors per chip 250 M gates per SOC are feasible by 2005, most SOCs would use only 50 M gates More design starts for Application-Specific Standard Products (ASSP) than for ASIC Increase of the number of programmable processing units Decrease of the gate count for custom logic blocks The trend is expected to continue, yielding a “sea of processors” Many heterogeneous processors connected by a network-on-chip System-level design tools combined with use of off-the-shelf components are needed Application-Specific Instruction-set Processors (ASIP) from ASSP Extensible processor platform is state-of-the-art in ASIP technology

178 Open Issues and Alternatives
Optimisation and search through a large design space Right set of extensible instructions and its constraints Communication of many extensible SOC processors on chip A customised Network-on-Chip (NOC) Shift in SoC design distinction To the process of customising extensible processors To the software design expertise needed to program them “Structured” ASICs introduce pre-built blocks (logic, configurable memories, and test structures) Most of the metal layers predefined Customisation of the upper two or three metal layers All customers share the same prefabricated die

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